Transmission signal generation apparatus, transmission signal generation method, reception signal apparatus, and reception signal method

ABSTRACT

Multiple-input multiple-output (MIMO) orthogonal frequency division multiplexing (OFDM) communication is provided which allows high accuracy estimation of frequency offset, high accuracy estimation of a transmission path fluctuation and high accuracy synchronization/signal detection. Pilot symbol mapping is provided for forming pilot carriers by assigning orthogonal sequences to corresponding subcarriers among OFDM signals which are transmitted at the same time from respective antennas in the time domain. Even when pilot symbols are multiplexed among a plurality of channels (antennas), this allows frequency offset/phase noise to be estimated with high accuracy.

CROSS-REFERENCE TO RELATED PRIORITY APPLICATIONS

This is a continuation application of U.S. application Ser. No.14/067,737 filed Oct. 30, 2013, which is a continuation of U.S.application Ser. No. 13/604,531 filed Sep. 5, 2012, which is acontinuation application of U.S. application Ser. No. 13/171,121 filedJun. 28, 2011, which is a continuation application of U.S. applicationSer. No. 12/840,024 filed Jul. 20, 2010, which is a divisionalapplication of U.S. application Ser. No. 11/577,791 filed Apr. 23, 2007,which is a national stage of PCT/JP2006/316653 filed Aug. 24, 2006,which is based on Japanese Application No. 2005-243494 filed Aug. 24,2005, and Japanese Application No. 2006-228337 filed Aug. 24, 2006, theentire contents of each of which are incorporated by reference herein.

TECHNICAL FIELD

The present invention relates to a Multiple-Input and Multiple-Output(MIMO)-Orthogonal Frequency Division Multiplexing (OFDM) transmissionapparatus and a MIMO-OFDM transmission method. More particularly, thepresent invention relates to a technology for realizing an ideal symbolconfiguration for frequency offset estimation, transmission pathfluctuation (channel fluctuation) estimation and synchronization/signaldetection, in MIMO-OFDM communication.

BACKGROUND

FIG. 1 shows the configuration of a transmission/reception apparatus ofa wireless LAN (Local Area. Network) which is an example of a radiocommunication system using OFDM (Orthogonal Frequency DivisionMultiplexing) which is currently being implemented, and the frameconfiguration thereof.

FIG. 1(a) shows an example of the configuration of a transmissionapparatus a id frame configuration signal generation section 10 receivescontrol information 9 such as a modulation scheme as input, determines aframe configuration a id outputs frame configuration signal 11.Serial/parallel conversion section (S/P) 2 receives frame configurationsignal 11 and baseband signal 1 subjected to digital modulation asinput, applies a serial/parallel conversion and outputs parallel signal3 in accordance with the frame configuration. Inverse Fourier transform(ifft) section 4 receives parallel signal 3 as input, applies InverseFourier transform and outputs signal 5 after the Inverse Fouriertransform. Radio section 6 receives signal 5 after the Inverse Fouriertransform as input, applies a frequency conversion or the like andoutputs transmission signal 7. Transmission signal 7 is transmitted as aradio wave from antenna 8.

FIG. 1(b) shows a configuration example of a reception apparatus andradio section 14 receives received signal 13 received at antenna 12 asinput, applies processing such as a frequency conversion and outputsbaseband signal 15. Synchronization section 16 receives baseband signal15 as input, establishes time synchronization with a transmitter andoutputs timing signal 17. Fourier transform (fft) section 18 receivesbaseband signal 15 and timing signal 17 as input, applies a Fouriertransform to baseband signal 15 based on timing signal 17 and outputssignal 19 after the Fourier transform.

Transmission path fluctuation estimation section 20 receives signal 19after the Fourier transform and timing signal 17 as input, detects apreamble in the signal after the Fourier transform, estimates atransmission path fluctuation and outputs transmission path fluctuationestimation signal 21. Frequency offset estimation section 22 receivessignal 19 after the Fourier transform and timing signal 17 as input,detects preambles and pilot symbols in the signal after the Fouriertransform, estimates frequency offset based on these symbols and outputsfrequency offset estimation signal 23.

Demodulation section 24 receives signal 19 after the Fourier transform,timing signal 17, transmission path fluctuation estimation signal 21 andfrequency offset estimation signal 23 as input, compensates for thetransmission path fluctuation and frequency offset in signal 19 afterthe Fourier transform, demodulates signal 19 and outputs receiveddigital signal 25.

FIG. 1(c) shows an image in a frame configuration of IEEE802.11a (not anexact frame configuration). The vertical axis shows frequency and thehorizontal axis shows time, and preambles are inserted at the head toestimate (detect a signal in some cases) a transmission path fluctuationand frequency offset. Furthermore, pilot symbols are inserted inspecific carriers such as carrier 2 and carrier 5 and used for thereceiver to estimate frequency offset and/or phase noise. The preamblesand pilot symbols are those whose signal point constellations on thein-phase I-quadrature Q plane are known. On the other hand, data istransmitted by means of data symbols.

The wireless LAN scheme is described in Non-Patent Document 1.

Non-Patent Document 1: “High speed physical layer (PHY) in 5 GHz band”IEEE802.11a, 1999

SUMMARY

Non-Patent Document 1 shows the symbol configuration for frequencyoffset estimation, transmission path fluctuation (channel fluctuation)estimation and synchronization/signal detection in a case using OFDM.

By the way, since further improvement of transmission speed can beexpected in the wireless LAN by combining the scheme disclosed inNon-Patent Document 1 with a MIMO system using spatial multiplexing orSDM: Spatial Division Multiplexing, it is possible to provide users awide variety of services.

Obtaining high reception quality in this MIMO-OFDM system requires highaccuracy frequency offset estimation, high accuracy transmission pathfluctuation estimation and high accuracy synchronization/signaldetection.

However, in the present circumstances, sufficient consideration has notbeen given to the method of transmitting symbols for transmission pathestimation and symbols for frequency offset estimation to realize highaccuracy frequency offset estimation, high accuracy transmission pathfluctuation estimation and high accuracy synchronization/signaldetection.

It is an object of the present invention to provide a MIMO-OFDMtransmission apparatus and a MIMO-OFDM transmission method capable ofhigh accuracy frequency offset estimation, high accuracy transmissionpath fluctuation estimation and high accuracy synchronization/signaldetection.

The MIMO-OFDM transmission apparatus according to the present inventiontransmits OFDM-modulated data symbols from a plurality of antennas in adata transmission period and transmits pilot symbols from specificcarriers of the plurality of antennas in the data transmission period,and employs a configuration having: an OFDM signal forming section thatforms OFDM signals transmitted from the antennas; and a pilot symbolmapping section that assigns orthogonal sequences to the same carriersin OFDM signals transmitted from the antennas at the same tit e, in thetime domain, and forms pilot carriers.

According to this configuration, orthogonal sequences are assigned tocorresponding subcarriers among OFDM signals transmitted at the sametime from the respective antennas in the time domain to form pilotcarriers, so that, even when pilot symbols are multiplexed among aplurality of channels (antennas), it is possible to estimate frequencyoffset/phase noise with high accuracy. Furthermore, since pilot symbolsof each channel can be extracted without using a channel estimationvalue (transmission path fluctuation estimation value), it is possibleto simplify the configuration of the section for compensating for thefrequency offset/phase noise.

Furthermore, the MIMO-OFDM transmission apparatus of the presentinvention adopts a configuration where, when the OFDM signals aretransmitted from two antennas, a pilot symbol mapping section formspilot carriers such that: pilot signals of orthogonal sequences are usedfor the same carriers between a first antenna and a second antenna;pilot signals of different sequences are used for different carriers atthe first antenna and the second antenna; and pilot signals or the samesequence are used at the first antenna and the second antenna.

According to this configuration, when MIMO-OFDM transmission is carriedout using two transmission antennas, it is possible to realize atransmission apparatus capable of minimizing an increase of transmissionpeak power in a simple configuration without degrading estimationaccuracy for frequency offset/phase noise.

Moreover, the MIMO-OFDM transmission apparatus of the present inventionadopts a configuration, wherein, when the OFDM signals are transmittedfrom three antennas, a pilot symbol mapping section forms pilot carrierssuch that: pilot signals of orthogonal sequences are used for the samecarriers among a first antenna, a second antenna and a third antenna;there is an antenna where pilot signals of different sequences are usedfor carriers assigned the pilot signals; and there are two or moreantennas where pilot signals of the same sequence are used.

According to this configuration, when MIMO-OFDM transmission is carriedout using three transmission antennas, it is possible to realize atransmission apparatus capable of minimizing an increase of transmissionpeak power in a simple configuration without degrading the estimationaccuracy for frequency offset/phase noise.

According to the present invention, it is possible to realize aMIMO-OFDM transmission apparatus and a MIMO-OFDM transmission methodcapable of high accuracy frequency offset estimation, high accuracytransmission path fluctuation estimation and high accuracysynchronization/signal detection.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 provides block diagrams illustrating a conventional radiocommunication system, where (a) shows a configuration example of atransmission apparatus, (b) shows a configuration example of a receptionapparatus and (c) shows a configuration example of transmission frame;

FIG. 2 is a block diagram showing the configuration of a MIMO-OFDMtransmission apparatus according to Embodiment 1 of the presentinvention;

FIG. 3A is a block diagram showing the configuration of the MIMO-OFDMreception apparatus according to Embodiment 1 of the present invention;

FIG. 3B shows a relationship between transmission and reception antennasaccording to Embodiment 1;

FIG. 4 provides diagrams showing frame a frame configuration of a signaltransmitted from each antenna of Embodiment 1, where (a) shows the frameconfiguration of channel A and (b) shows the frame configuration ofchannel B;

FIG. 5 provides diagrams showing a signal point constellation of datasymbols, where FIG. 5A shows a signal point constellation of BPSK, FIG.5B shows a signal point constellation of QPSK, FIG. 5C shows a signalpoint constellation of 16QAM, FIG. 5D shows a signal point constellationof 64QAM and FIG. 5E shows normalization factors by which signals of therespective modulation schemes are multiplied;

FIG. 6 illustrates signal point constellations of pilot symbolsaccording to Embodiment 1;

FIG. 7 is a block diagram showing the configuration of frequencyoffset/phase noise compensation section;

FIG. 8 illustrates signal point constellations of preambles according toEmbodiment 1;

FIG. 9 is a block diagram showing the configuration of a transmissionpath fluctuation estimation section;

FIG. 10 illustrates signal point constellations of preambles accordingto Embodiment 1;

FIG. 11 shows time fluctuation of reception intensity of preambles anddata symbols;

FIG. 12 illustrates signal point constellations of preambles accordingto Embodiment 1;

FIG. 13 is a block diagram showing the configuration of a mappingsection according to an embodiment of the present invention;

FIG. 14 is a block diagram showing the configuration of a MIMO-OFDMtransmission apparatus according to Embodiment 2;

FIG. 15 is a block diagram showing the configuration of a MIMO-OFDMreception apparatus according to Embodiment 2;

FIG. 16 shows a relationship between transmission and reception antennasaccording to Embodiment 2;

FIG. 17 provides diagrams showing a frame configuration of a signaltransmitted from each antenna according to Embodiment 2, where (a) showsthe frame configuration of channel A, (b) shows the frame configurationof channel B and (c) shows the frame configuration of channel C;

FIG. 18 illustrates signal point constellations of preambles accordingto Embodiment 2;

FIG. 19 illustrates signal point constellations of preambles accordingto Embodiment 2;

FIG. 20 provides block diagrams showing the configuration of Embodiment3, where (a) shows the configuration of a terminal and (b) shows a blockdiagram of the configuration of an access point;

FIG. 21 shows a relationship between a communication scheme and anormalization factor according to Embodiment 3;

FIG. 22 illustrates signal point constellations of preambles accordingto Embodiment 3;

FIG. 23 illustrates signal point constellations of preambles ofaccording to Embodiment 3;

FIG. 24 provides diagrams showing the frame configuration according toEmbodiment 4, where (a) shows the frame configuration of channel A, (b)shows the frame configuration of channel B and (c) shows the frameconfiguration of channel C;

FIG. 25 shows an example of the configuration of preambles according toEmbodiment 4;

FIG. 26 shows another equation of a relationship between a communicationscheme and a normalization factor;

FIG. 27 provides diagrams showing the frame configuration according toEmbodiment 5, where (a) shows the frame configuration of channel A, (b)shows the frame configuration of channel B and (c) shows the frameconfiguration of channel C;

FIG. 28 illustrates signal point constellations of pilot symbolsaccording to Embodiment 5;

FIG. 29 is a block diagram showing the configuration of a MIMO-OFDMtransmission apparatus according to Embodiment 5;

FIG. 30 is a block diagram showing the configuration of a mappingsection according to Embodiment 5;

FIG. 31 is a block diagram showing another example of the configurationof a mapping section according to Embodiment 5;

FIG. 32 is a block diagram showing the configuration of frequencyoffset/phase noise compensation section according to Embodiment 5;

FIG. 33 is a block diagram showing another configuration of thefrequency offset/phase noise compensation section according toEmbodiment 5;

FIG. 34 provides diagrams showing other examples of the frameconfiguration according to Embodiment 5, where (a) shows the frameconfiguration of channel A, (b) shows the frame configuration of channelB and (c) shows the frame configuration of channel C;

FIG. 35 provides diagrams showing further examples of the frameconfiguration according to Embodiment 5, where (a) shows the frameconfiguration of channel A, (b) shows the frame configuration of channelB and (c) shows the frame configuration of channel C;

FIG. 36 provides diagrams showing still further examples of the frameconfiguration according to Embodiment 5, where (a) shows the frameconfiguration of channel A, (b) shows the frame configuration of channelB and (c) shows the frame configuration of channel C;

FIG. 37 is a block diagram showing the configuration of frequencyoffset/phase noise estimation section according to Embodiment 6;

FIG. 38 is a block diagram showing the configuration of a mappingsection according to Embodiment 6;

FIG. 39 provides diagrams showing the frame configuration according toEmbodiment 7, where (a) shows the frame configuration of channel A and(b) shows the frame configuration of channel B;

FIG. 40 illustrates signal point constellations of pilot symbolsaccording to Embodiment 7;

FIG. 41 is a block diagram showing the configuration of a MIMO-OFDMtransmission apparatus according to Embodiment 7;

FIG. 42 is a block diagram showing the configuration of a mappingsection according to Embodiment 7;

FIG. 43 is a block diagram showing the configuration of frequencyoffset/phase noise estimation section according to Embodiment 7;

FIG. 44 is a block diagram showing the configuration of a MIMO-OFDMtransmission apparatus according to Embodiment 8;

FIG. 45 provides diagrams showing the frame configuration according toEmbodiment 8, where (a) shows the frame configuration of channel A, (b)shows the frame configuration of channel B, (c) shows the frameconfiguration of channel C and (d) shows the frame configuration ofchannel D;

FIG. 46 shows a relationship between a reference symbol modulationscheme and a normalization factor when carrying out transmission basedon a quadruple transmission spatial multiplexing MIMO scheme;

FIG. 47 shows mapping examples of reference symbols when carrying outtransmission based on a quadruple transmission spatial multiplexing MIMOscheme;

FIG. 48 shows mapping examples of reference symbols when carrying outtransmission based on a quadruple transmission spatial multiplexing MIMOscheme; and

FIG. 49 provides diagrams showing other examples of the frameconfiguration according to Embodiment 8, where (a) shows the frameconfiguration of channel A, (b) shows the frame configuration of channelB, (c) shows the frame configuration of channel C and (d) shows theframe configuration of channel D.

DESCRIPTION OF EXAMPLE EMBODIMENTS

Hereinafter, embodiments of the present invention will be explained indetail with reference to the attached drawings.

Embodiment 1

This embodiment will explain the configuration of a MIMO system usingspatial multiplexing and the configurations of a transmission apparatusand a reception apparatus in the system, and will also explain theconfigurations of pilot symbols, preambles and reference symbols whichenables the improvement of estimation accuracy of frequency offset,transmission path fluctuation and synchronization, and the probabilityof signal detection.

FIG. 2 shows the configuration of MIMO-OFDM transmission apparatus 100according to this embodiment. However, FIG. 2 shows a case where thenumber of transmission antennas m=2 as an example.

Frame configuration signal generation section 112 receives controlinformation 111 on a modulation scheme or the like as input, generatesframe configuration signal 113 which includes information on the frameconfiguration and outputs this.

Mapping section 102A receives transmission digital signal 101A ofchannel A and frame configuration signal 113 as input, generatesbaseband signal 103A based on the frame configuration and outputs this.

Serial/parallel conversion section 104A receives baseband signal 103Aand frame configuration signal 113 as input, applies a serial/parallelconversion thereto based on frame configuration signal 113 and outputsparallel signal 105A.

Inverse Fourier transform section 106A receives parallel signal 105A asinput, applies an inverse Fourier transform thereto and outputs signal107A after the inverse Fourier transform.

Radio section 108A receives signal 107A after the inverse Fouriertransform as input, applies processing such as a frequency conversionand outputs transmission signal 109A. Transmission signal 109A is outputas a radio wave from antenna 110A.

MIMO-OFDM transmission apparatus 100 also generates transmission signal109B of channel B by applying processing similar to that on channel A tochannel B. The element indicated with “B” appended at the end of thereference numeral is a part related to channel B, which simply meansthat the target signal is not channel A but channel B and is basicallysubjected to processing similar to that on the above described partrelated to channel A indicated with “A” appended at the end of thereference numeral.

FIG. 3A shows an example of the configuration of a reception apparatusaccording to this embodiment. However. FIG. 3A shows a case where thenumber of reception antennas 2 as an example.

In reception apparatus 200, radio section 203X receives signal 202Xreceived at reception antenna 201X as input, applies processing such asa frequency conversion thereto and outputs baseband signal 204X.

Fourier transform section 205X receives baseband signal 204X as input,applies a Fourier transform and outputs signal 206X after the Fouriertransform.

A similar operation is, also carried out on the reception antenna 201Yside and synchronization section 211 receives baseband signals 204X and204Y as input, establishes time synchronization with a transmitter bydetecting, for example, reference symbols and outputs timing signal 212.The configuration or the like of reference symbols will be explained indetail using FIG. 4 or the like.

Frequency offset/phase noise estimation section 213 receives signals206X and 206Y after the Fourier transform as input, extracts pilotsymbols and estimates frequency offset and/or phase noise from the pilotsymbols and outputs phase distortion estimation signal 214 (phasedistortion including frequency offset). The configuration or the like ofpilot symbols will be explained in detail using FIG. 4 or the like.

Transmission path fluctuation estimation section 207A of channel Areceives signal 206X after the Fourier transform as input, extractsreference symbols of channel A, estimates a transmission pathfluctuation of channel A based on the reference symbols, for example,and outputs transmission path estimation signal 208A of channel A.

Transmission path fluctuation estimation section 207B of channel. Breceives signal 206X after the Fourier transform as input, extractsreference symbols of channel B, estimates a transmission pathfluctuation of channel B based on the reference symbols, for example,and outputs transmission path estimation signal 208B of channel B.

Transmission path fluctuation estimation section 209A of channel A andtransmission path fluctuation estimation section 209B of channel Breceive a signal received at antenna 201Y as the target signal insteadof a signal received at antenna 201X and basically carry out processingsimilar to that described above on transmission path fluctuationestimation section 207A of channel A and transmission path fluctuationestimation section 20713 of channel B.

Frequency offset/phase noise compensation section 215 receivestransmission path estimation signals 208A and 210A of channel A,transmission path estimation signals 208B and 210B of channel B, signals206X and 206Y after the Fourier transform and phase distortionestimation signal 214 as input, removes the phase distortion of eachsignal and outputs transmission path estimation signals 220A and 222A ofchannel A after phase compensation, transmission path estimation signals220B and 222B of channel B after phase compensation and signals 221X and221Y after the Fourier transform and after phase compensation.

Signal processing section 223 carries out, for example, an inversematrix operation and outputs baseband signal 224A of channel A andbaseband signal 224B of channel B. More specifically, as shown in FIG.3B, for example, assuming that for a certain subcarrier, a transmissionsignal from antenna AN1 is Txa(t), a transmission signal from antennaAN2 is Txb(t), a received signal of antenna AN3 is R1(t), a receivedsignal of antenna AN4 is R2(t) and transmission path fluctuations areh11(t), h12(t), h21(t) and h22(t) respectively, the followingrelationship equation holds.

$\begin{matrix}{\left\lbrack {{Equation}\mspace{14mu} 1} \right\rbrack\mspace{619mu}} & \; \\{\begin{pmatrix}{R\; 1(t)} \\{R\; 2(t)}\end{pmatrix} = {{\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}{{Txa}(t)} \\{{Txb}(t)}\end{pmatrix}} + \begin{pmatrix}{n\; 1(t)} \\{n\; 2(t)}\end{pmatrix}}} & (1)\end{matrix}$

where t is time, n1(t) and n2(t) is noise. Signal processing section 223obtains a signal of channel A and a signal of channel B by carrying out,for example, an operation of an inverse matrix using Equation (1).Signal processing section 223 executes this operation on allsubcarriers. h11(t), h12(t), h21(t) and h22(t) are estimated bytransmission path fluctuation estimation sections 207A, 209A, 207B and209B.

Frequency offset estimation/compensation section 225A receives basebandsignal 224A of channel A as input, extracts pilot symbols, estimates andcompensates for frequency offset of baseband signal 224A based on thepilot symbols and outputs baseband signal 226A after frequency offsetcompensation.

Channel A demodulation section 227A receives baseband signal 226A afterthe frequency offset compensation as input, demodulates data symbols andoutputs received data 228A.

MIMO-OFDM reception apparatus 200 also applies similar processing tobaseband signal 224B of channel B and obtains received data 228B.

FIG. 4 shows a time-frequency frame configuration of channel A (FIG.4(a)) and channel B (FIG. 4(b)) of this embodiment. Signals at the sametime on the same carriers in FIG. 4(a) and FIG. 4(b) are spatiallymultiplexed.

From time 1 to time 8, symbols for estimating transmission pathfluctuations corresponding to h11(t), h12(t), h21(t) and 1)220) inEquation (1), for example, symbols called “preambles” are transmitted.This preamble is composed of guard symbol 301 and reference symbol 302.Guard symbol 301 is assumed to be (0, 0) on the in-phase I-quadrature Qplane. Reference symbol 302 is, for example, a symbol at knowncoordinates other than (0, 0) on the in-phase I-quadrature Q plane.Channel A and channel B are configured such that no interference occurswith each other. That is, when, for example, guard symbol 301 is placedon channel A on carrier 1 at time 1, reference symbol 302 is placed onchannel B and when reference symbol 302 is placed on channel A oncarrier 2 at time 1, guard symbol 301 is placed on channel B, and inthis way different symbols are placed on channel A and channel B. Byadopting such an arrangement, when, for example, attention is focused onchannel A at time 1, it is possible to estimate a transmission pathfluctuation of carrier 3 using reference symbols 302 of carrier 2 andcarrier 4. Since carrier 2 and carrier 4 are reference symbols 302, thetransmission path fluctuation can be estimated. Therefore, transmissionpath fluctuations of all carriers of channel A can be accuratelyestimated at time 1. In the same way, transmission path fluctuations ofall carriers can be accurately estimated for channel B, too. From time 2to time 8, transmission path fluctuations of all carriers of channel Aand channel B can be estimated in the same way. Thus, since the frameconfiguration in FIG. 4 allows transmission path fluctuations of allcarriers to be estimated at all times from time 1 to time 8, this can besaid to be a preamble configuration capable of realizing quite highaccuracy estimation of transmission path fluctuations.

In FIG. 4, information symbol (data symbol) 303 is a symbol fortransmitting data. Here, suppose the modulation scheme is BPSK, QPSK,16QAM or 64QAM. A signal point constellation on the in-phaseI-quadrature Q plane or the like in this case will be explained indetail using FIG. 5.

Control symbol 304 is a symbol for transmitting control information on amodulation scheme, error correcting coding scheme, coding rate or thelike.

Pilot symbol 305 is a symbol for estimating a phase fluctuation byfrequency offset and/or phase noise. For example, a symbol having knowncoordinates on the in-phase I-quadrature Q plane is used as pilot symbol305. Pilot symbols 305 are placed on carrier 4 and carrier 9 on bothchannel A and channel B. When a wireless LAN builds a system at the samefrequency and in the same frequency band according to IEEE802.11a,IEEE802.11g and spatial multiplexing MIMO system, this allows the frameconfiguration to be shared, and therefore it is possible to simplify thereception apparatus.

FIG. 5 shows signal point constellations of BPSK, QPSK, 16QAM and 64QAMwhich are modulation schemes of information symbols 303 in FIG. 4 on thein-phase I-quadrature Q plane and normalization factors thereof.

FIG. 5A shows a signal point constellation of BPSK on the in-phaseI-quadrature Q plane and their coordinates are as shown in FIG. 5A. FIG.5B is a signal point constellation of QPSK on the in-phase I-quadratureQ plane and their coordinates are as shown in FIG. 5B. FIG. 5C is asignal point constellation of 16QAM on the in-phase I-quadrature; Qplane and their coordinates are as shown FIG. 5D is a signal pointconstellation of 64QAM on the in-phase I-quadrature Q plane and theircoordinates are as shown in FIG. 5D. FIG. 5E shows a relationshipbetween a modulation scheme and a multiplication factor (i.e.,normalization factor) for correcting signal point constellations in FIG.5A to FIG. 5D so that average transmit power is kept constant amongdifferent modulation schemes. For example, when transmission is carriedout under the modulation scheme of QPSK in FIG. 5B, as is clear fromFIG. 5E, it is necessary to multiply the coordinates in FIG. 5B by avalue of 1/sqrt(2). Here, sqrt(x) refers to the square root of x.

FIG. 6 shows an arrangement of pilot symbols 305 in FIG. 4 on thein-phase I-quadrature Q plane according to this embodiment. FIG. 6(a)shows an example of signal point constellation of pilot symbols 305 fromtime 11 to time 18 on carrier 4 of channel A shown in FIG. 4(a). FIG.6(b) shows an example of signal point constellation of pilot symbols 305from time 11 to time 18 on carrier 4 of channel B shown in FIG. 4(b).FIG. 6(e) shows an example of signal point constellation of pilotsymbols 305 from time 11 to time 18 on carrier 9 of channel A shown inFIG. 4(a). FIG. 6(d) shows an example of signal point constellation ofpilot symbols 305 from time 11 to time 18 on carrier 9 of channel Bshown in FIG. 4(b). Here, BPSK modulation is used for these arrangementsbut the modulation is not limited to this.

A feature of the signal point constellations of pilot symbols 305 inFIG. 6 is that the signal point constellations on the same carriers ofchannel A and channel B are orthogonal to each other (cross-correlationis zero).

For example, the signal point constellation from time 11 to time 14 ofchannel A on carrier 4 is orthogonal to the signal point constellationfrom time 11 to time 14 of channel B on carrier 4. Furthermore, the sameapplies to time 15 to time 18. The signal point constellation from time11 to time 14 of channel A on carrier 9 is also orthogonal to the signalpoint constellation from time 11 to time 14 of channel B on carrier 9.Furthermore, the same applies to time 15 to time 18. At this time, usinga Walsh-Hadamard conversion, orthogonal codes or the like is suitablebecause of the orthogonality of signals. FIG. 6 shows the case of BPSK,but if signals are orthogonal, QPSK modulation may also be used or itmay not be necessary to follow the rule of the modulation scheme.

Furthermore, in the case of this embodiment, for simplicity of thereceiver, suppose the signal point constellations are the same (samepattern) between carrier 4 of channel A and carrier 9 of channel B andbetween carrier 9 of channel A and carrier 4 of channel B (here, asshown in FIG. 6, they are named “pattern #1” and “pattern #2”). Thereason will be explained in detail in FIG. 7. However, the same patterndoes not mean to adopt completely the same signal point constellation.For example, a case where only the phase relationship is different onthe in-phase I-quadrature Q plane can also be regarded as the samepattern.

Furthermore, the signal point constellation of pilot symbols 305 is madeto differ between carriers 4 and 9 of channel A (or channel B) and thisis because using the same signal point constellation may lead to anincrease of transmission peak power. However, the pattern defined abovemay be the same. In other words, it is important that the signal pointconstellations are different.

Here, an advantage of orthogonality will be explained in detail usingFIG. 3A and FIG. 7 first.

FIG. 7 is an example of the configuration of frequency offset/phasenoise estimation section 213 in FIG. 3A. Pilot carrier extractionsection 602 receives signal 206X (or 206Y) after a Fourier transform asinput and extracts subcarrier which are pilot symbols 305. Morespecifically, it extracts signals of carrier 4 and carrier 9. Therefore,pilot carrier extraction section 602 outputs baseband signal 603 ofcarrier 4 and baseband signal 604 of carrier 9.

Code storage section 605 stores, for example, pattern #1 in FIG. 6 andoutputs signal 606 of pattern #1 according to timing signal 212.

Code storage section 607 stores, for example, pattern #2 in FIG. 6 andoutputs signal 608 of pattern #2 according to timing signal 212.

Selection section 609 receives timing signal 212, signal 606 of pattern#1 and signal 608 of pattern #2 as input, outputs the signal of pattern#2 as selection signal 610(X) and outputs the signal of pattern #1 asselection signal 611(Y).

Code multiplication section 612A receives baseband signal 603 of carrier4 and selection signal 611(Y) as input, multiplies baseband signal 603of carrier 4 by selection signal 611(Y), generates baseband signal 613Aof channel A of carrier 4 and outputs this to phase fluctuationestimation section 616A. The reason is as follows.

Baseband signal 603 of carrier 4 is a signal in which the basebandsignal of channel A and the baseband signal of channel B aremultiplexed. On the other hand, when this is multiplied by selectionsignal 611(Y), that is, the signal of pattern #1, the component of thebaseband signal of channel B whose cross-correlation is zero is removedand only the component of the baseband signal of channel A can beextracted.

In the same way, code multiplication section 614A receives basebandsignal 604 of carrier 9 and selection signal 610(X) as input, multipliesbaseband signal 604 of carrier 9 by selection signal 610(X), generatesbaseband signal 615A of channel A of carrier 9 and outputs this to phasefluctuation estimation section 618A.

Code multiplication section 612B receives baseband signal 603 of carrier4 and selection signal 610(X) as input, multiplies baseband signal 603of carrier 4 by selection signal 610(X), generates baseband signal 613Bof channel B of carrier 4 and outputs this to phase fluctuationestimation section 616B.

Code multiplication section 614B receives baseband signal 604 of carrier9 and selection signal 611(Y) as input, multiplies baseband signal 604of carrier 9 by selection signal 611 (Y), generates baseband signal 615Bof channel B of carrier 9 and outputs this to phase fluctuationestimation section 618B.

As described above, by making signal point constellations of channel Aand channel B on the same carriers orthogonal to each other, even whenpilot symbols 305 are multiplexed on channel A and channel B, it ispossible to estimate frequency offset and/or phase noise with highaccuracy. Another important advantage is that since no channelestimation value (transmission path fluctuation estimation value) isrequired, it is possible to simplify the configuration of the part forcompensating for the frequency offset and/or phase noise. If signalpoint constellations of pilot symbols 305 of channel A and channel B arenot orthogonal to each other, signal processing of MIMO demultiplexing,for example, ZF (Zero Forcing), MMSE (Minimum Mean Square Error) or MLD(Maximum Likelihood Detection) is carried out, frequency offset and/orphase noise are then estimated. On the other hand, according to theconfiguration of this embodiment, it is possible to compensate forfrequency offset and/or phase noise before demultiplexing a signal(signal processing section 223) as shown in FIG. 3A. In addition, thefrequency offset and/or phase noise can be removed using pilot symbols305 even after demultiplexing the signal of channel A from the signal ofchannel B by signal processing section 223, thereby making it possibleto compensate for the frequency offset and/or phase noise with higheraccuracy.

By the way, when the signal, point constellations of channel A andchannel B of the same carriers are not orthogonal to each other, theestimation accuracy for frequency offset and/or phase noise by frequencyoffset/phase noise estimation section 213 in FIG. 3A decreases (signalsbecome components of interference with each o(her), and therefore it isdifficult to add the frequency offset/phase noise compensation section215 in FIG. 3A and it is not possible to realize high accuracy frequencyoffset/phase noise compensation.

Furthermore, this embodiment assumes the same signal point constellation(same pattern) for carrier 4 of channel A and carrier 9 of channel B,and carrier 9 of channel A and carrier 4 of channel B, thereby providingcommonality between code storage sections 605 and 607 in FIG. 7 andleading to simplification of the reception apparatus.

However, while it is essential in this embodiment that signal pointconstellations of channel A and channel B on the same carriers beorthogonal to each other, it is not necessarily essential that they havethe same pattern.

This embodiment has been explained using an example where pilot symbols305, which are orthogonal to each other in four-symbol units, from time11 to time 14, but the present invention is not limited to four-symbolunits. However, when considering an influence on the orthogonality dueto a fluctuation of fading in the time direction, if an orthogonalpattern is formed in 2 to 8-symbol units, may be possible to secure theestimation accuracy for frequency offset/phase noise. When the period ofan orthogonal pattern is too long, the possibility that theorthogonality may not be secured increases and the estimation accuracyfor frequency offset/phase noise degrades. Furthermore, the case wherethe number of transmission antennas is 2 and two modulated signals aretransmitted has been explained, but the present invention is not limitedto this and even in a case where the number of transmission antennas is3 or more and three or more modulated signals are transmitted, it ispossible to obtain effects similar to those described above by makingpilot symbols 305 existing on the same carriers orthogonal to each otherin several-symbol units.

Next, the configuration of reference symbols 302 in preambles of FIG. 4which makes it possible to simplify the reception apparatus and suppressthe degradation of the accuracy of transmission path estimation due toquantization errors which occur at the reception apparatus will beexplained in detail.

FIG. 8 shows signal point constellations of preambles on the in-phaseI-quadrature Q plane according to this embodiment, especially signalpoint constellations at times 1, 3, 5, 7 at which reference symbols 302are arranged on carriers 2, 4, 6, 8, 10, 12.

Here, signals formed at times 1, 3, 5, 7 of carrier 2, signals formed attimes 1, 3, 5, 7 of carrier 4, signals formed at times 1, 3, 5, 7 ofcarrier 6, signals formed at times 1, 3, 5, 7 of carrier 8, signalsformed at times 1, 3, 5, 7 of carrier 10 and signals formed at times 1,3, 5, 7 of carrier 12 have the same pattern on the in-phase I-quadratureQ plane though their phase relationships are different. This cansimplify the reception apparatus.

Likewise adopting the same pattern for carriers 1, 3, 5, 7, 9, 11, tooleads to simplification of the reception apparatus though their phaserelationships are different. Here, making the pattern of even numberedcarriers and the pattern of the odd numbered carrier same can furthersimplify the reception apparatus. However, even when they are different,there is an advantage to a certain degree in the aspect ofsimplification of the reception apparatus. This is because only onenecessary pattern signal is added. Similarly, adopting the same patternfor channel A and channel B leads to further simplification of thereception apparatus, but adopting different patterns also has anadvantage to a certain degree.

Hereinafter, the point related to the simplification of theconfiguration of the reception apparatus will be explained. However, acase where the pattern of even numbered carriers and the pattern of oddnumbered carriers are the same will be explained as an example.

FIG. 9 shows the configuration of the details of transmission pathfluctuation estimation sections 207 and 209 of the reception apparatusin FIG. 3A. Here, estimation of the transmission path fluctuation ofchannel A will be explained as an example.

Signal extraction section 802_1 of carrier 1 receives signal 206X (206Y)after a Fourier transform as input, extracts signals corresponding toreference symbols 302 (times 2, 4, 6, 8) of carrier 1 in the preamblesof channel A shown in FIG. 4(a) and outputs reference signal 803_1 ofcarrier 1.

Signal extraction section 802_2 of carrier 2 receives signal 206X (206Y)after a Fourier transform as input, extracts signal corresponding toreference symbols 302 (times 1, 3, 5, 7) of carrier 2 in the preamblesof channel A shown in FIG. 4(a) and outputs reference signal 803_2 ofcarrier 2.

Similar operations are also performed at signal extraction sections ofcarrier 3 to carrier 12.

Pattern signal generation section 804 outputs pattern signal 805 of (1,0), (−1, 0), (−1, 0), (1, 0) on the in-phase I-quadrature Q plane (seethe pattern in FIG. 8).

Multiplication section 806_1 receives reference signal 803_1 of carrier1 and pattern signal 805 as input, multiplies reference signal 803_1 ofcarrier 1 by pattern signal 805 and applies signal processing ofaveraging or the like and outputs transmission path fluctuationestimation signal 807_1 of carrier 1 to transmission path fluctuationcalculation section 808_1.

Multiplication section 806_2 to multiplication section 806_12 alsooperate in the same way and output transmission path fluctuationestimation signal 807_2 of carrier 2 to transmission path fluctuationestimation signal 807_12 of carrier 12 to transmission path fluctuationcalculation section 808_2 through 808_12.

Parallel/serial conversion section 810 receives transmission pathfluctuation estimation signal 807_1 to transmission path fluctuationestimation signal 807_12 of carrier 1 to carrier 12 as input, applies aparallel/serial conversion and outputs transmission path fluctuationestimation signal 208A (208B, 210A, 210B).

In this way, pattern signal generation section 804 can be shared amongcarrier 1 to carrier 12, and therefore it is possible to reduce thestorage capacity of pattern signals of pattern signal generation section804 and standardize the signal processing and simplify the receptionapparatus accordingly.

By the way, what is shown in FIG. 8 are signal point constellations onthe in-phase I-quadrature Q plane when reference symbols 302 aremodulated in BPSK and they are similar to the signal pointconstellations when data symbols 303 are modulated in BPSK and thenormalization factors to be multiplied are also similar to those whendata symbols 303 are modulated in BPSK. However, by so doing, if ananalog/digital converter is mounted to carry out digital signalprocessing at the reception apparatus, the influence of quantizationerrors increases. An example of signal point constellation on thein-phase I-quadrature Q plane to solve this problem will be explained.

FIG. 10 shows an example of signal point constellation on the in-phaseI-quadrature Q plane for solving this problem. As an example. BPSKmodulation is used. At this time, suppose that the normalization factoris 1.0 and the signal point constellation of reference symbols 302 is(1.414≈sqrt(2), 0) or (−1.414.≈sqrt(2), 0). In other words, a signalpoint constellation is assumed to be obtained by multiplying the signalpoint constellation when data symbols 303 are modulated in BPSK bycoefficient 1.414.

Time fluctuation in the intensity of the received signal in such a casewill be explained using FIG. 11.

In FIG. 11, FIG. 11(a) shows time fluctuation of the waveform of thereceived signal when the signal point constellation of preambles in FIG.8 is adopted and FIG. 11(b) shows time fluctuation of the waveform ofthe received signal when the signal point constellation of preambles inFIG. 10 is adopted. When the signal point constellation of preambles asshown in FIG. 8 is adopted, average reception power of preambles issmaller than average reception power of data symbols 303. Thisphenomenon is caused by the existence of guard symbols 301 when the samesignal point constellation as that of data symbols 303 is carried outfor reference symbols 302 of preambles. As a result, especially when areceived signal is converted to a digital signal using an analog/digitalconverter, the quality of the received signals of preambles degrades dueto the influence of quantization errors.

On the other hand, when the signal point constellation of preambles asshown in FIG. 10 is carried out, as shown in FIG. 11(b), the averagereception power of preambles becomes equivalent to the average receptionpower of data symbols 303. Therefore, even if a received signal isconverted to a digital signal using an analog/digital converter, theinfluence of quantization errors on the received signals of preambles isreduced and the quality is secured.

Based on a concept similar to that described above, FIG. 12 shows amethod of signal point constellation when QPSK is adopted as the signalpoint constellation of reference symbols 302.

FIG. 12 show an example of signal point constellation on the in-phaseI-quadrature Q plane when QPSK modulation is applied to referencesymbols 302 assuming that a normalization factor is 1. In this way, asshown in FIG. 11(b), the average reception power of preambles isequivalent to the average reception power of data symbols 303 and evenif a signal is converted to a digital signal using an analog/digitalconverter, the influence of quantization errors on the received signalsof preambles is reduced and the quality is secured.

As described above, when the modulation scheme called #X of data symbols303 is used for reference symbols 302, it is important to use the signalpoint constellation which becomes 1.414.≈.sqrt(2) times the signal pointconstellation of data symbols 303 on the in-phase I-quadrature Q planeafter multiplication by a normalization factor as the signal pointconstellation of reference symbols 302 on the in-phase I-quadrature Qplane after multiplication by a normalization factor. The factor1.414.≈.sqrt(2) is a value determined because reference symbols 302 arearranged for every symbol on the frequency axis.

For example, when #X is QPSK, the signal point constellation on thein-phase I-quadrature Q plane after multiplication by a normalizationfactor is (.±1/sqrt(2), ±1/sqrt(2)) and the signal point constellationof reference symbols 302 on the in-phase I-quadrature Q plane aftermultiplication by a normalization factor becomes (±1, ±1) according tothe above described rule (see FIG. 12).

FIG. 13 shows an example of the configuration of mapping section 102A(102B) of the transmission apparatus of this embodiment in FIG. 2. Datamodulation section 1103 receives transmission digital signal 101A (101B)and frame configuration signal 1102 as input, applies modulation totransmission digital signal 101A (101B) based on the information on themodulation scheme and timing included in frame configuration signal 1102and outputs modulated signal 1104 of data symbols 303.

Preamble mapping section 1105 receives frame configuration signal 1102as input and outputs modulated signal 1106 of preambles based on theframe configuration.

Code storage section #1 (1107) outputs signal 1108 of pattern #1. In thesame way, code storage section #2 (1109) outputs signal 1110 of pattern#2.

Pilot symbol mapping section 1111 receives signal 1108 of pattern #1,signal 1110 of pattern #2 and frame configuration signal 1102 as input,generates modulated signal 1112 of pilot symbol 305 and outputs this.

Signal generation section 1113 receives modulated signal 1104 of datasymbols 303, modulated signal 1106 of preambles and modulated signal1112 of pilot symbols 305 as input, generates baseband signal 103A(103B) in accordance with the frame configuration and outputs this.

The above described explanation shows that adopting the configuration ofpilot symbols 305 as shown in FIG. 4 and FIG. 6 can simplify thereception apparatus. Similarly, adopting the pilot symbol configurationas shown in FIG. 4 and FIG. 6 also allows the transmission apparatus tostandardize code storage sections 1107 and 1109 as shown in FIG. 13 andalso leads to the simplification of the transmission apparatus.

As shown above, the method of generating preambles and pilot signals(pilot symbols) of this embodiment, the detailed configuration andoperation of the transmission apparatus which generates them and thereception apparatus which receives the modulated signal of thisembodiment have been explained. According to this embodiment, it ispossible to improve the estimation accuracy of frequency offset,transmission path fluctuation and synchronization, thereby improving theprobability of detection of signals and simplifying the transmissionapparatus and the reception apparatus.

In other words, an important feature of the above described embodimentis a MIMO-OFDM transmission apparatus that transmits OFDM-modulated datasymbols 303 from a plurality of antennas for a data transmission periodand transmits OFDM-modulated symbols for transmission path estimationfrom the above described plurality of antennas for a period differentfrom the above described data transmission period including a datamapping section (data modulation section 1103) that forms data symbols303, a symbol mapping section for transmission path estimation (preamblemapping section 1105) that forms symbols for transmission pathestimation for which, when assuming that the number of subcarriers isin, the signal point amplitude of n subcarriers is 0, α=m/(m−n), thesignal point amplitude of the remaining m-n subcarriers becomes is timesthe signal point amplitude based on the same modulation scheme among themodulation schemes of data symbols 303 and an OFDM modulation sectionthat OFDM-modulates above described data symbols 303 and the abovedescribed symbols for transmission path estimation. This makes itpossible to reduce quantization errors of symbols for transmission pathestimation on the receiving side, thereby realizing transmission pathfluctuation estimation with high accuracy.

This embodiment has been explained with an example using an OFDM scheme,but the present invention is not limited to this and can also beimplemented in the same way when using a single-carrier scheme, othermulticarrier schemes or a spread spectrum communication scheme.Furthermore, this embodiment has been explained with an example wheretwo antennas are used for transmission and reception respectively, butthe present invention is not limited to this and even when the number ofreception antennas is three or more, this embodiment will not beinfluenced and can be implemented in the same way. Furthermore, theframe configuration is not limited to this embodiment and especially,pilot symbols 305 for estimating distortion such as frequency offsetand/or phase noise are only required to be arranged on specificsubcarriers and transmitted from a plurality of antennas and the numberof subcarriers which transmit pilot symbols 305 is not limited to two inthis embodiment. Embodiments with other numbers of antennas or usingother transmission methods will be explained in detail later. Inaddition, this embodiment has been explained using naming such as “pilotsymbol 305”, “reference symbol 302”, “guard symbol 301”, “preamble”, butusing other names will by no means influence this embodiment. This willbe the same in other embodiments.

Embodiment 2

This embodiment will explain in detail a case where the number oftransmission/reception antennas in Embodiment 1 is assumed to be three.

FIG. 14 shows an example of the configuration of a transmissionapparatus according to this embodiment. In FIG. 14, components whichoperate in the same way as those in FIG. 2 are assigned the samereference numerals as those in FIG. 2. MIMO-OFDM transmission apparatus1200 in FIG. 14 is different from FIG. 2 in that a transmission sectionof channel C is added thereto.

FIG. 15 shows an example of the configuration of a reception apparatusaccording to this embodiment. In FIG. 15, components which operate inthe same way as those in FIG. 3 are assigned the same referencenumerals. In FIG. 15, since modulated signals of three channels aretransmitted from the transmission apparatus, transmission pathfluctuation estimation sections 207C and 209C of channel C are added andone antenna is added compared to the configuration in FIG. 3A andnecessary configurations corresponding thereto are added.

FIG. 16 shows a relationship between transmission and reception antennasin this embodiment. For example, the following relationship equationholds if it is assumed in a certain subcarrier that a transmissionsignal from antenna 1401 is Txa(t), transmission signal from antenna1402 is Txb(t), transmission signal from antenna 1403 is Txc(t),received signal of antenna 1404 is R1(t), received signal of antenna1405 is R2(t), received signal of antenna 1406 is R3(t) and theirrespective transmission path fluctuations are h11(t), h12(t), h13(t),h21(t), h22(t), h23(t), h31(t), h32(t) and h33(t).

$\begin{matrix}{\left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack\mspace{619mu}} & \; \\{\begin{pmatrix}{R\; 1(t)} \\{R\; 2(t)} \\{R\; 3(t)}\end{pmatrix} = {{\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} & {h\; 13(t)} \\{h\; 21(t)} & {h\; 22(t)} & {h\; 23(t)} \\{h\; 31(t)} & {h\; 32(t)} & {h\; 33(t)}\end{pmatrix}\begin{pmatrix}{{Txa}(t)} \\{{Txb}(t)} \\{{Txc}(t)}\end{pmatrix}} + \begin{pmatrix}{n\; 1(t)} \\{n\; 2(t)} \\{n\; 3(t)}\end{pmatrix}}} & (2)\end{matrix}$

where t is time and n1(t), n2(t) and n3(t) are noise. Signal processingsection 223 in FIG. 15 carries out, for example, an operation of aninverse matrix using Equation (2) to thereby obtain a signal of channelA, a signal of channel B and a signal of channel C. Signal processingsection 223 carries out this operation on all subcarriers. h11(t),h12(t), h13(t), h21(t), h22(t), h23(t), h31(t), h32(t) and h33(t) areestimated by transmission path fluctuation estimation sections 207A,209A, 1301A, 20713, 209B, 1301B, 207C, 209C and 1301C.

FIG. 17 shows an example of the frame configuration according to thisembodiment and parts corresponding to those in FIG. 4 are assigned thesame reference numerals. FIG. 17(a) shows the time-frequency frameconfiguration of channel A, FIG. 17(b) shows the time-frequency frameconfiguration of channel B and FIG. 17(c) shows an example of thetime-frequency frame configuration of channel C. Signals at the sametime and on the same carrier of channels A, 13 and C in FIG. 17(a), FIG.17(b) and FIG. 17(c) are spatially multiplexed.

From time 1 to time 8, symbols for estimating transmission pathfluctuations corresponding to h11(t), h12(t), h13(t), h21(t), h22(t),h23(t), h31(t), h32(t) and h33(t) in Equation (2) are transmitted. Thissymbol is composed of guard symbol 301 and reference symbol 302. Supposeguard symbol 301 is (0, 0) on the in-phase I-quadrature Q plane.Reference symbol 302 is, for example, a symbol at known coordinatesother than (0, 0) on the in-phase I-quadrature Q plane. Channel A,channel B and channel C are configured in such a way that nointerference occurs with each other. That is, when, for example,reference symbol 302 is placed on channel A on carrier 1 at time 1,guard symbol 301 is placed on channel B and channel C, when referencesymbol 302 is placed on channel B on carrier 2 at time 1, guard symbol301 is placed on channel A and channel C, and when reference symbol 302is placed on channel C on carrier 3 at time 1, guard symbol 301 isplaced on channel A and channel B. In this way, reference symbol 302 isplaced on only one channel on a certain carrier at a certain time andguard symbols 301 are placed on other channels. By adopting suchplacement, when, for example, attention is focused of channel A at time1, transmission path fluctuations of carriers 2 and 3 can be estimatedusing reference symbols 302 on carrier 1 and carrier 4. Since carrier 1and carrier 4 are reference symbols 302, transmission path fluctuationscan be estimated. Therefore, transmission path fluctuations of allcarriers of channels A can be accurately estimated at time 1. In thesame way, transmission path fluctuations of all carriers of channel Band channel C can be accurately estimated, too. From time 2 to time 8,the transmission path fluctuations of all carriers of channel A, channelB and channel C can be estimated in the same way. Thus, since thetransmission path fluctuations of all carriers can be estimated at alltimes from time 1 to time 8, the frame configuration in FIG. 17 can besaid to be a preamble configuration capable of realizing estimation oftransmission path fluctuations with quite high accuracy.

Next, when all analog/digital converter is mounted for the receptionapparatus to perform digital signal processing, a signal pointconstellation of preambles (especially reference symbols 302) on thein-phase I-quadrature plane for reducing the influence of quantizationerrors will be explained. As a precondition, suppose that the signalpoint constellations of data symbols 303 in FIG. 17 are adopted and thenormalization factors according to FIG. 5 are adopted.

FIG. 18 shows an example (constellation examples at times 1, 2, 3 ofchannel A) of signal point constellation of preambles on the in-phaseI-quadrature Q plane. Here, suppose the modulation scheme of referencesymbols 302 is BPSK. At this time, a case where a normalization factoris 1 and the signal point constellations in FIG. 5A are adopted isassumed, the reception intensity of preambles decreases compared to datasymbols 303 as shown in (a) of FIG. 11, and therefore the influence ofquantization errors increases on only preambles due to theanalog/digital converter mounted on the reception apparatus, causingdeterioration of the reception quality. On the other hand, in FIG. 18,the signal point constellation of reference symbol 302 on the in-phaseI-quadrature Q plane is assumed to be (1.732≈sqrt(3), 0) or(−1.732.≈sqrt(3), 0). That is, the signal point constellationcorresponds to a signal point constellation, of data symbols based on aBPSK modulation multiplied by factor 1.732.

The image of intensity of the received signals of preambles and datasymbols 303 on the time axis is as shown in FIG. 11(b). This allows theinfluence of quantization errors on preambles by the analog/digitalconverter mounted on the reception apparatus to be reduced, andtherefore the reception quality improves.

Based on the above described concept, FIG. 19 shows the method of signalpoint constellation when QPSK is assumed to be the signal pointconstellation of reference symbols 302.

FIG. 19 shows an example of signal point constellation on the in-phaseI-quadrature Q plane when QPSK modulation is applied to referencesymbols 302 assuming that a normalization factor is 1. In this way, asshown in FIG. 11(b), the average reception power of preambles isequivalent to the average reception power of data symbols 303 and evenif a signal is converted to a digital signal using an analog/digitalconverter, the influence of quantization errors on the received signalsof preambles is reduced and the quality is secured. Here, 1.225 in FIG.19 is calculated from 1.255≈(3)/sqrt(2).

As described above, when the modulation scheme called #X of data symbols303 is used for reference symbols 302, it is important to use the signalpoint constellation which becomes 1.732≈sqrt(3) times the signal pointconstellation of data symbol 303 on the in-phase I-quadrature Q planeafter multiplication by a normalization factor as the signal pointconstellation of reference symbols 302 on the in-phase I-quadrature Qplane after multiplication by a normalization factor. The factor1.732≈sqrt(3) is a value determined because reference symbols 302 arearranged for every two symbols on the frequency axis.

For example, when #X is QPSK, the signal point constellation on thein-phase I-quadrature Q plane after multiplication by a normalizationfactor is (±1/sqrt(2), ±1/sqrt(2)) and the signal point constellation ofreference symbols 302 on the in-phase I-quadrature Q plane aftermultiplication by a normalization factor becomes (±sqrt(3)/sqrt(2),±sqrt(3)/sqrt(2)) according to the above described role (see FIG. 19).

As shown above, the method of generating preambles and pilot signals ofthis embodiment, the detailed configuration and operation of thetransmission apparatus which generates them and the reception apparatuswhich receives a modulated signal of this embodiment, especially in thecase where the number of the transmission/reception antennas is three,have been explained. According to this embodiment, it is possible toimprove the accuracy of estimation of frequency offset, transmissionpath fluctuation and synchronization, thereby improving the probabilityof detection of signals and simplifying the transmission apparatus andthe reception apparatus.

This embodiment has been explained with an example using an OFDM scheme,but the present invention is not limited to this and can also beimplemented in the same way when using single-carrier schemes, othermulticarrier schemes or spread spectrum communication schemes.Furthermore, even when the number of reception antennas is four or more,this embodiment will not be influenced and can be implemented in thesame way.

Embodiment 3

This embodiment will explain details of the configuration of preamblesin a communication scheme whereby a spatial multiplexing MIMO system inwhich the number of transmission antennas is two and the number oftransmission modulated signals is two (double transmission spatialmultiplexing MIMO) and a spatial multiplexing MIMO system in which thenumber of transmission antennas is three and the number of transmissionmodulated signals is three (triple transmission spatial multiplexingMIMO) are switched round according to a communication environment (e.g.,reception quality or the like).

FIG. 20 shows a communication mode of this embodiment. FIG. 20(a) showsa terminal and FIG. 20(b) shows an access point (AP). At this time, acase where the AP changes a MIMO scheme and a modulation method will beexplained as an example.

In the terminal in FIG. 20(a), transmission apparatus 1902 receivestransmission digital signal 1901 as input, outputs modulated signal 1903and modulated signal 1903 is output as a radio wave from antenna 1904.At this time, suppose that transmission digital signal 1901 includesinformation on communication situation for the AP to change acommunication scheme, for example, a bit error rate, packet error rate,reception field intensity or the like.

In the AP of FIG. 20(b), reception apparatus 1907 receives signal 1906received at antenna 1905 as input and outputs received digital signal1908.

Transmission method determining section 1909 receives received digitalsignal 1908 as input, determines a communication method (that is, a MIMOscheme and a modulation scheme) based on information on a communicationsituation included in received digital signal 1908 and outputs controlinformation 1910 including this information.

Transmission apparatus 1912 receives control information 1910 andtransmission digital signal 1911 as input, modulates transmissiondigital signal 1911 based on the determined communication method,outputs modulated signal 1913 and this modulated signal 1913 istransmitted front the antenna.

FIG. 14 shows an example of the detailed configuration of transmissionapparatus 1912 in FIG. 20(b). Frame configuration signal generationsection 112 in FIG. 14 receives control information 111—that is, controlinformation 1910 in FIG. 20—as input, determines a modulation scheme anda MIMO scheme based on this and outputs frame configuration signal 113including this information. For example, the transmission section ofchannel C does not operate when frame configuration signal 113 indicatesa double transmission spatial multiplexing MIMO scheme. This allows thedouble transmission spatial multiplexing MIMO scheme and the tripletransmission spatial multiplexing MIMO scheme to be switched round.

Here, the frame configuration when a double transmission spatialmultiplexing MIMO scheme is selected becomes as shown in FIG. 4explained in Embodiment 1 and the frame configuration when a tripletransmission spatial multiplexing MIMO scheme is selected becomes asshown in FIG. 17 explained in Embodiment 2.

As for the signal point constellation on the in-phase I-quadrature Qplane of data symbols 303, those shown in FIG. 5A, FIG. 5B, FIG. 5C andFIG. 5D are adopted. However, as for the normalization factors, thoseshown in FIG. 21 are adopted.

FIG. 21 shows a normalization factor to be adopted in each MIMO schemeand each modulation scheme when a normalization factor is assumed to be1 in the case of a double transmission spatial multiplexing MIMO andBPSK. The reason that normalization factors are set in this way is toequalize total transmit power of modulated signals transmitted by the APregardless of the modulation scheme and regardless of the number ofmodulated signals to be transmitted. Therefore, if the normalizationfactor based on the double transmission spatial multiplexing MIMO schemeis assumed to be X when the same modulation Scheme is used, thenormalization factor of the triple transmission spatial multiplexingMIMO scheme becomes sqrt(2)/sqrt(3) times X.

Next, a signal point constellation of preambles on the in-phaseI-quadrature Q plane at this time will be explained.

In the case of the double transmission spatial multiplexing MIMO scheme,suppose that the signal point constellation of reference symbols 302 ofpreambles on the in-phase I-quadrature Q plane is as shown in FIG. 10 orFIG. 12 explained in Embodiment 1 in order to reduce the influence ofquantization errors produced in the analog/digital converter of thereception apparatus.

Considering the normalization factors in FIG. 21 and the explanations ofEmbodiment 2 at this time, it is necessary to adopt signal pointconstellations as shown in FIG. 22 and FIG. 23 to reduce the influenceof quantization errors produced in the analog/digital converter of thereception apparatus. By so doing, even when the AP switches between thedouble transmission spatial multiplexing MIMO scheme and the tripletransmission spatial multiplexing MIMO scheme, the reception apparatuscan reduce the influence of quantization errors on preambles.

What is important in the above described explanation, when the totaltransmit power of modulated signals to be transmitted is equalizedregardless of the modulation scheme and the number of modulated signalsto be transmitted as shown in FIG. 21 is that when using a modulationscheme called “#X” of data symbols 303 for reference symbols 302, thesame signal point constellation of reference symbols 302 on the in-phaseI-quadrature Q plane after multiplication by a normalization factor beused for the double transmission spatial multiplexing MIMO scheme andthe triple transmission spatial multiplexing MIMO scheme.

Adopting such a configuration can reduce the influence of quantizationerrors in preambles, thereby suppressing degradation of receptionquality.

The case where the average reception power of data symbols 303 isequalized to that of the preambles has been explained above as anexample, but it is often the case that reception quality is secured byusing greater average reception power of preambles than the averagereception power of data symbols 303. The above-described concept canalso be applied to this case, too. In short, when using the modulationscheme called #X of data symbols 303 for reference symbols 302, it isonly necessary to observe the rule that the same signal pointconstellation of reference symbols 302 on the in-phase I-quadrature Qplane after multiplication by a normalization factor is used for thedouble transmission spatial multiplexing MIMO scheme and the tripletransmission spatial multiplexing MIMO scheme.

That is, in an n-transmission spatial multiplexing MIMO scheme having ntransmission antennas and n transmission modulated signals, whenreference symbol 302 is inserted for every n−1 symbols at the preamble,if a modulation scheme called #X of data symbols 303 is used forreference symbols 302, the operation would be the same as that describedabove if the same signal point constellation of reference symbols 302 onthe in-phase I quadrature Q plane after multiplication by anormalization factor is used regardless of n.

This embodiment has been explained with an example using an OFDM scheme,but the present invention is not limited to this and can also beimplemented in the same way when using a single-carrier scheme, othermulticarrier schemes or spread spectrum communication schemes.Furthermore, even when the number of reception antennas is three ormore, this embodiment will not be thereby influenced and can beimplemented in the same way.

Embodiment 4

Embodiment 2 has explained the configuration of preambles of a tripletransmission spatial multiplexing MIMO system. When using preambleconfigurations like Embodiments 1 and 2, the interval at which referencesymbols 302 exist increases as the number of antennas increases, andtherefore the possibility that the estimation accuracy of transmissionpath fluctuations at the reception apparatus may degrade increases. Thisembodiment proposes a method for preamble configuration to diminish thisp

FIG. 24 shows an example of frame configuration according to thisembodiment. A characteristic part in F16.24 is the configuration ofpreambles. The basic operation when transmitting the frame signal inFIG. 24 is the same as that in the case of transmitting the frame signalin FIG. 17 explained in Embodiment 2 and signals of channel A, channel Band channel C on the same carrier at the same time are transmitted fromdifferent antennas and spatially multiplexed.

in FIG. 24, at time 1, all carriers 1 to 12 of channel A are composed ofreference symbols 302. Then, at time 2, all carriers 1 to 12 of channelB are composed of reference symbols 302. At time 3, all carriers 1 to 12of channel C are composed of reference symbol 302.

FIG. 25 shows a detailed configuration of preambles of carrier 1 andcarrier 2 at time 1 and time 2 in particular. At time 1, a generalOFDM-based modulated signal is generated on channel A whose all carriers1 to 12 are reference symbols 302. Channel B and channel C on whichreference symbols 302 and guard symbols 301 exist in carriers 1 to 12are subjected to special signal processing after Inverse Fouriertransformer 106 in FIG. 14. In the same way, general OFDM-basedmodulated signals are generated on channel B at time 2, on channel C attime 3, on channel A at time 4, . . . , and so on. Special signalprocessing is applied after Inverse Fourier transformer 106 in FIG. 14to channel A and channel C at time 2, channel A and channel B at time 3,channel B and channel C at time 4, and so on.

In FIG. 25, suppose a signal point constellation of (I, Q)=(1, 0) on thein-phase I-quadrature Q plane is performed on channel A at time 1 oncarrier 1. Interference with other channels is prevented by setting thesignal point constellation of channel C to (I, Q)=(0, 0) at this time.Then, suppose that the same signal point constellation as that forchannel A is used for channel B. The above described special signalprocessing is then applied thereto. This special signal processing isthe processing on channel B for shifting the phase of a modulated signalobtained after a Fourier transform, for example, by 0.5 symbols withrespect to the time axis. However, signals which stick out of channel Awithout overlapping channel A are placed (turned down) one by one fromtop to bottom. This operation is disclosed in the document “Channelestimation for OFDM systems with transmitter diversity in mobilewireless channels,” IEEE Journal on Selected Areas in Communications,vol. 17, no. 3, pp-461-471, March 1999 and “Simplified channelestimation for OFDM systems with multiple transmit antennas,” IEEETransaction on wireless communications, vol. 1, no. 1, pp. 67-75, 2002,and Cyclic delay diversity is applied to channel A and channel B at time1 on carrier 1.

In the same way, suppose that a signal point constellation of (I,Q)=(−1, 0) is performed on channel A at time 1 on carrier 2 on thein-phase I-quadrature Q plane. Interference with other channels isprevented by setting the signal point constellation of channel B to (I,Q)=(0, 0) at this time. Then, suppose that channel C is subjected to thesame signal point constellation as that for channel A. The abovedescribed special signal processing is then applied thereto. Thisspecial signal processing is the processing on channel C for shiftingthe phase of a modulated signal obtained after a Fourier transform, forexample, by 0.5 symbols with respect to the time axis. However, signalswhich stick out of channel A without overlapping channel A are placed(turned down) one by one from top to bottom. Therefore, cyclic delaydiversity is applied to channel A and channel C at time 1 on carrier 2.

Therefore, at time 1, there is a situation in which cyclic delaydiversity is applied to channel A and channel B, and channel A andchannel C alternately with respect to the frequency axis. At this time,the transmission path fluctuation estimation section of the receptionapparatus can estimate transmission paths of channels to which cyclicdelay diversity is applied by realizing equalization processing.Therefore, it is possible to estimate transmission path fluctuations ofchannel A and channel B at the same time on carrier 1 at time 1 andestimate transmission path fluctuations of channel A and channel C oncarrier 2 at time 1. Unlike the case of Embodiment 2, this allowstransmission path fluctuations corresponding to two channels to beestimated at the same time, and therefore the number of preamble symbolscan be reduced. This leads to improvement of the data transmissionspeed.

However, while cyclic delay diversity capable of estimating transmissionpath fluctuations corresponding to three channels is theoreticallypossible, there is a disadvantage that the diversity gain decreases andthe circuit scale of the reception apparatus increases, and thereforethe configuration whereby transmission path fluctuations correspondingto two channels are estimated at the same time like this embodiment ispreferable. For this purpose, it is important to always insert guardsymbol ((I, Q)=(0, 0)) in any one of channels.

When attention is focused on time 1, the phases of channel B and channelC are shifted, and it is very important to match the amount of phaseshifted (may also be expressed by the amount of symbols or time), forexample, 0.5 symbols. This is because the reception apparatus canthereby standardize the circuit of simultaneous estimation oftransmission path fluctuations of channel A and channel B andsimultaneous estimation of transmission path fluctuations of channel Aand channel C.

As shown above, arranging preambles of triple transmission spatialmultiplexing MIME) system so that cyclic delay diversity of two channelsis executed on a channel which differs from one carrier to another makesit possible to estimate transmission path fluctuations with highaccuracy and also reduce the number of preambles and improve the datatransmission speed.

This embodiment has explained the case of triple transmission, but canalso be applied to a case with four or more antennas. This embodimenthas been explained with an example using an OFDM scheme, but the presentinvention is not limited to this and can also be implemented in the sameway when using a single-carrier scheme, other multicarrier schemes or aspread spectrum communication scheme

Embodiment 3 above has explained switching between double transmissionspatial multiplexing MIMO and triple transmission spatial multiplexingMIMO, but the present invention is not limited to this. The presentinvention can be likewise implemented even when, for example, switchinginvolves single transmission (when MIMO is not executed). Therelationship between a communication scheme and a normalization factorin that case becomes as shown in FIG. 26.

Embodiment 5

While Embodiment 1 has explained the case with two transmissionantennas, this embodiment will explain a pilot carrier configurationwhen the number of the transmission antennas is three.

FIG. 27 shows an example of the frame configuration of a transmissionsignal which is formed by a transmission apparatus of this embodimentand components corresponding to those in FIG. 4 are assigned the samereference numerals. FIG. 27(a) shows the frame configuration of channelA, FIG. 27(b) shows the frame configuration of channel B and FIG. 27(c)shows the frame configuration of channel C. As is clear from FIG. 27,pilot symbols (pilot carriers) 305 are arranged on carrier 3, carrier 5,carrier 8, carrier 10 except for times at which reference symbols 302and control symbols 304 are transmitted.

FIG. 28 shows signal point constellations of pilot symbols 305 ofchannel A, channel B and channel C, and a feature thereof. The featurein this case is that signal point constellations of channel A, channel Band channel C on the same carriers are orthogonal to each other(cross-correlation is zero) as in the case of Embodiment 1.

For example, the signal point constellation (FIG. 28(a)) from time 11 totime 14 of channel A on carrier 3, the signal point constellation (FIG.28(b)) from time 11 to time 14 of channel B on carrier 3 and the signalpoint constellation (FIG. 28(c)) from time 11 to time 14 of channel C oncarrier 3 are orthogonal to each other. Signal point constellations areperformed so that such orthogonality is also maintained from time 15onward. Using Walsh-Hadamard conversion, orthogonal codes or the like issuitable because of the orthogonality of signals at this time. FIG. 28shows a case of BPSK, but if signals are orthogonal, QPSK modulation mayalso be used or it may be possible not to follow the rule of themodulation scheme.

Furthermore, for simplicity of a transmission apparatus and a receptionapparatus in the case of this embodiment, the same signal pointconstellation (same sequence) is used between carrier 3 of channel A(FIG. 28(a)) and carrier 5 of channel C (FIG. 28(f)), between carrier 3of channel B (FIG. 28(b)) and carrier 8 of channel B (FIG. 28(h)),between carrier 3 of channel C (FIG. 28(c)) and carrier 10 of channel A(FIG. 28(j)), between carrier 5 of channel A (FIG. 28(d)) and carrier 8of channel C (FIG. 28(i)), between carrier 5 of channel B (FIG. 28(e))and carrier 10 of channel B (FIG. 28(k)), and between carrier 8 ofchannel A (FIG. 28(g)) and carrier 10 of channel C (FIG. 28(l)). Thereason will be explained in detail using FIG. 30, FIG. 31, FIG. 32 andFIG. 33. Here, the “same sequence” means completely the same signalpoint constellation. Here, the respective sequences are named sequence#1, sequence #2, sequence #3, sequence #4, sequence #5, sequence #6 asshown in FIG. 28.

Furthermore, different signal point constellations (different sequences)of pilot symbols 305 are used for carriers 3, 5, 8, 10 of channel A (orchannel C) and this is because there is a possibility that using thesame signal point constellation (same sequence) may lead to an increaseof transmission peak power. However, channel B in this embodiment is anexample of not meeting this condition.

Here, the simplification of the transmission apparatus and the receptionapparatus and the necessity for orthogonality will be explained indetail using FIG. 29, FIG. 30, FIG. 31, FIG. 32 and FIG. 33.

FIG. 29 shows an example of the configuration of a MIMO-OFDMtransmission apparatus according to this embodiment. In FIG. 29, partsthat operate in the same way as those in FIG. 14 are assigned the samereference numerals as those in FIG. 14. In MIMO-OFDM transmissionapparatus 2800, mapping section 2802 receives transmission data 2801 andframe configuration signal 113 as input and outputs baseband signal 103Aof channel A, baseband signal 103B of channel B and baseband signal 103Cof channel C. The other parts operate in a manner similar to thatexplained in Embodiment 1 or Embodiment 2, and therefore explanationsthereof will be omitted.

FIG. 30 shows an example of the detailed configuration of mappingsection 2802 in FIG. 29. Data modulation section 2902 receivestransmission data 2801 and frame configuration signal 113 as input,generates modulated signal 2903 of data symbols 303 according to frameconfiguration signal 113 and outputs this.

Preamble mapping section 2904 receives frame configuration signal 113 asinput, generates modulated signal 2905 of preambles according to frameconfiguration signal 113 and outputs this.

Sequence #1 storage section 2906 outputs signal 2907 of sequence #1 inFIG. 28. Sequence #2 storage section 2908 outputs signal 2909 ofsequence #2 in FIG. 28. Sequence #3 storage section 2910 outputs signal2911 of sequence #3 in FIG. 28. Sequence #4 storage section 2912 outputssignal 2913 of sequence #4 in FIG. 28. Sequence #5 storage section 2914outputs signal 2915 of sequence #5 in FIG. 28. Sequence #6 storagesection 2916 outputs signal 2917 of sequence #6 in FIG. 28.

Pilot symbol mapping section 2918 receives signal 2907 of sequence #1,signal 2909 of sequence #2, signal 2911 of sequence #3, signal 2913 ofsequence #4, signal 2915 of sequence #5, signal 2917 of sequence #6 andframe configuration signal 113 as input, generates a modulated signal ofpilot symbols 305 according to frame configuration signal 113 andoutputs this.

Signal generation section 2920 receives modulated signal 2903 of datasymbols 303, modulated signal 2905 of preambles and modulated signal2919 of pilot symbols 305 as input and outputs modulated signal 103A ofchannel A, modulated signal 103B of channel B and modulated signal 103Cof channel C.

The configuration in FIG. 30 requires only six sequence storagesections. This is because as shown in FIG. 28, the present inventionuses a certain sequence in two or more subcarriers (used in twosubcarriers in FIG. 28). This allows the circuit scale of thetransmission apparatus to be reduced. On the other hand, unlike FIG. 28,using all different sequences would require 12 sequence storagesections, which would increase the circuit scale.

Mapping section 2802 in FIG. 29 may also be composed, for example, asshown in FIG. 31. In FIG. 31, components which operate in the same wayas those in FIG. 30 are assigned the same reference numerals as those inFIG. 30. Code #1 storage section 3001 stores “1, 1, −1, −1” and code #2storage section 3003 stores “1, −1, 1, −1”. Pilot symbol mapping section2919 receives signal 3002 of pattern #1 and signal 3004 of pattern #2output from code #1 storage section 3001 and code #2 storage section3003, and frame configuration signal 113 as input and outputs modulatedsignal 2920 of pilot symbols 305.

At this time, as is clear from FIG. 28, there are only two types ofbasic signal pattern. Pilot symbol mapping section 2919 shifts codesusing a shift register so as to generate six types of sequences #1 to #6from the two basic patterns. Therefore, as shown in FIG. 31, the storagesection can be composed of only two sections.

As described above, as is understandable from FIG. 29, FIG. 30 and FIG.31, configuring pilot carriers as shown in FIG. 28 allows theconfiguration of the transmission apparatus to be simplified.

Next, the reception apparatus will be explained. FIG. 15 is an exampleof the configuration of the reception apparatus. Hereinafter, theconfiguration of frequency offset/phase noise estimation section 213 inFIG. 15 will be explained in detail using FIG. 32 and FIG. 33.

FIG. 32 is an example of the configuration of frequency offset/phasenoise estimation section 213 in FIG. 15 according to this embodiment.Frequency offset/phase noise estimation section 213 in FIG. 32 isprovided with pilot carrier extraction section 3101, sequence storagesections 3108_1 to 3108_6, sequence selection section 3110, frequencyoffset/phase noise estimation section 3123_#3 of carrier 3, frequencyoffset/phase noise estimation section 3123_#5 of carrier 5, frequencyoffset/phase noise estimation section 3123_#8 of carrier 8 and frequencyoffset/phase noise estimation section 3123_#10 of carrier 10.

Pilot subcarrier extraction section 3101 receives signal 206X (or 206Y,206Z) after a Fourier transform as input and extracts subcarriers whichare pilot symbols 305. More specifically, signals of carriers 3, 5, 8,10 are extracted. Therefore, pilot subcarrier extraction section 3101outputs baseband signal 3102_#3 of carrier 3, baseband signal 3102_#5 ofcarrier 5, baseband signal 3102_#8 of carrier 8 and baseband signal3102_#10 of carrier 10.

Sequence #1 storage section 3108_1 stores sequence #1 in FIG. 28 andoutputs signal 3109_1 of sequence #1 according to timing signal 212.Sequence #2 storage section 3108_2 stores sequence #2 in FIG. 28 andoutputs signal 3109_2 of segue according to timing signal 212. Sequence#3 storage section 3108_3 stores sequence #3 in FIG. 28 and outputssignal 3109_3 of sequence #3 according to timing signal 212. Sequence #4storage section 3108_4 stores sequence #4 in FIG. 28 and outputs signal3109_4 of sequence #4 according to timing signal 212. Sequence #5storage section 3108_5 stores sequence #5 in FIG. 28 and outputs signal3109_5 of sequence #5 according to timing signal 212. Sequence #6storage section 3108_6 stores sequence #6 in FIG. 28 and outputs signal3109_6 of sequence #6 according to timing signal 212.

Sequence selection section 3110 receives signal 3109_1 of sequence #1,signal 3109_2 of sequence #2, signal 3109_3 of sequence #3, signal3109_4 of sequence #4, signal 3109_5 of sequence #5, signal 3109_6 ofsequence #6 and timing signal 212 as input and assigns sequence #5 tosignal 3111, sequence #1 to signal 3112, sequence #4 to signal 3113,sequence #3 to signal 3114, sequence #6 to signal 3115, sequence #5 tosignal 3116, sequence #2 to signal 3117, sequence #1 to signal 3118,sequence #3 to signal 3119, sequence #4 to signal 3120, sequence #6 tosignal 3121 and sequence #2 to signal 3122, and outputs them.

Frequency offset/phase noise estimation section 3123_#3 of carrier 3 isprovided with code multiplication sections 3103A, 3103B and 3103C, phasefluctuation estimation sections 3105A, 3105B and 3105C and estimatesfrequency offset and/or phase noise of each channel of carrier 3.

Code multiplication section 3103A receives baseband signal 3102_#3 ofcarrier 3 and signal 3111 of sequence #5 as input and multipliesbaseband signal 3102_#3 of carrier 3 by signal 3111 of sequence #5,thereby generating baseband signal 3104A_#3 of channel A of carrier 3and outputting this. The reason is as follows

Baseband signal 3102_#3 of carrier 3 is a signal in which the basebandsignal of channel A, the baseband signal of channel B and the basebandsignal of channel C are multiplexed. When this multiplexed signal ismultiplied by signal 3111 of sequence #5, the components of the basebandsignal of channel B and the baseband signal of channel C having a zerocross-correlation are removed and it is thereby possible to extract onlya baseband signal component of channel A.

Phase fluctuation estimation section 3105A receives baseband signal3104A_#3 of channel A of carrier 3 as input, estimates a phasefluctuation based on this signal and outputs phase fluctuationestimation signal 3106A_#3 of channel A.

In the same way, code multiplication section 3103B receives basebandsignal 3102_#3 of carrier 3 and signal 3112 of sequence #1 as input andmultiplies baseband signal 3102_#3 of carrier 3 by signal 3112 ofsequence #1, thereby generating baseband signal 3104B_#3 of channel B ofcarrier 3 and outputting this. Furthermore, code multiplication section3103C receives baseband signal 3102_#3 of carrier 3 and signal 3113 ofsequence #4 as input and multiplies baseband signal 3102_#3 of carrier 3by signal 3113 of sequence #4, thereby generating baseband signal3104C_#3 of channel C of carrier 3 and outputting this.

Phase fluctuation estimation sections 3105B and 3105 C receive basebandsignal 3104B_#3 of channel B of carrier 3 and baseband signal 3104C_#3of channel C of carrier 3 as input respectively, estimate phasefluctuations based on these signals and output phase fluctuationestimation signal 3106B_#3 of channel B and phase fluctuation estimationsignal 3106C_#3 of channel C respectively.

Frequency offset/phase noise estimation section 3123_#5 of carrier 5only differs in the signal to be processed and operates in the same wayas above described frequency offset/phase noise estimation section3123_#3 of carrier 3 and outputs phase fluctuation estimation signal3106A_#5 of channel A, phase fluctuation estimation signal 3106B_#5 ofchannel B and phase fluctuation estimation signal 3106C_#5 of channel Cabout carrier 5. Frequency offset/phase noise estimation section 3123_#8of carrier 8 only differs in the signal to be processed and operates inthe same way as above described frequency offset/phase noise estimationsection 3123_#3 of carrier 3 and outputs phase fluctuation estimationsignal 3106A_#8 of channel A, phase fluctuation estimation signal3106B_#8 of channel B and phase fluctuation estimation signal 3106C_#8of channel C about carrier 8. Moreover, frequency offset/phase noiseestimation section 3123_#10 of carrier 10 also only differs in thesignal to be processed and operates in the same way as above describedfrequency offset/phase noise estimation section 3123_#3 of carrier 3 andoutputs phase fluctuation estimation signal 3106A_#10 of channel A,phase fluctuation estimation signal 3106B_#10 of channel B and phasefluctuation estimation signal 3106C #10 of channel C about carrier 10.

As described above, causing signals of channel A, channel B and channelC of the same carrier to be orthogonal to each other, allows frequencyoffset/phase noise to be estimated with high accuracy even when pilotsymbols 305 are multiplexed on channel A, channel B and channel C.Another important advantage is that since no channel estimation value(transmission path fluctuation estimation value) is required, theconfiguration of the part for compensating for the frequencyoffset/phase noise can be simplified. If the signal point constellationsof pilot symbols 305 of channel A, channel B and channel C are notorthogonal to each other, signal processing of the MIMO demultiplexing,for example, ZF, MMSE, MILD are carried out and then frequency offsetand/or phase noise are estimated. On the other hand, according to thepresent invention, it is possible to compensate for the frequencyoffset/phase noise before demultiplexing a signal (signal processingsection 223) as shown in FIG. 15. In addition, signal processing section223 can remove the frequency offset/phase noise using pilot symbols 305even after the signal is demultiplexed into the signal of channel A,signal of channel B and signal of channel C, and therefore it ispossible to compensate for frequency offset and/or phase noise withhigher accuracy.

On the other hand, when the signal point constellations of channel A,channel B and channel C OD the same carriers are not orthogonal to eachother, the estimation accuracy for frequency offset/phase noiseestimation section 213 in FIG. 15 decreases (signals become componentsof interference with each other), and therefore it is difficult to addfrequency offset/phase noise compensation section 215 in FIG. 15 and itis not possible to perform frequency offset/phase noise compensationwith high accuracy.

Furthermore, as a configuration different from the configuration in FIG.32, the configuration in FIG. 33 may be considered. FIG. 33 is differentfrom

FIG. 32 in that sequence storage sections 3108_1 to 3108_6 are replacedby code storage sections 3201_1 and 3201_2. Code #1 storage section3201_1 stores “1, 1, −1, −1” and code 112 storage section 3201_2 stores“1, −1, 1 −1”. Code selection section 3203 shifts two types of basiccodes input from code storage sections 3201_1 and 3201_2 using a shiftregister, thereby generating six types of sequences #1 to #6. Thisallows the storage section to be composed of only two sections and canthereby simplify the configuration compared to the configuration in FIG.32. This is possible because pilot symbols 305 of the same sequence areassigned to a plurality of channels or a plurality of carriers.

As described above, a plurality of channels or a plurality of carriersuse pilot symbols 305 of the same sequence, and therefore it is possibleto standardize sequence storage sections 3108_1 to 3108_6 in FIG. 32 orstandardize code storage sections 3201_1 and 3201_2 in FIG. 33 among aplurality of channels or a plurality of carriers, which leads tosimplification of the reception apparatus.

As shown in FIG. 27, the case where pilot symbols (pilot carriers) 305for estimating distortion due to frequency offset or phase noise or thelike are arranged on specific subcarriers has been explained so far, butthe frame configuration of pilot symbols 305 different from FIG. 27 willbe explained hereinafter.

FIG. 34, FIG. 35 and FIG. 36 show examples of the frame configuration oftransmission signals different from that in FIG. 27.

In FIG. 34, pilot symbols 305 are arranged at specific times on specificcarriers. FIG. 34(a) shows the frame configuration of channel A, FIG.34(b) shows the frame configuration of channel B and FIG. 4(c) shows theframe configuration of channel C. In the example in FIG. 34, pilotsymbol sequences which are orthogonal to each other are used amongchannels of the same carriers and at the same time pilot symbolsequences of the same sequences are repeatedly used where pilot symbols305 are multiplexed on channel A, channel B and channel C. On channel A,pilot symbols 305 of different sequences are used for differentsubcarriers. That is, from time 6 to time 9 in the example of FIG. 34,mapping similar to that for pilots from time 11 to time 14 in FIG. 28 isperformed, and next from time 12 to time 15 in FIG. 34, mapping isperformed according to a rule similar to that from time 11 to time 14 inFIG. 28. Thus, using the frame configuration in FIG. 34 under acondition equivalent to the above described one can obtain effectssimilar to those described above.

In FIG. 35, pilot symbols 305 are arranged in a plurality of consecutivesubcarriers at a specific time. At this time, for example, a pilotsymbol sequence on channel A at time 6 on carrier 2 to carrier 5, apilot symbol sequence on channel B at time 6 on carrier 2 to carrier 5and a pilot symbol sequence on channel C at time 6 on carrier 2 tocarrier 5 are orthogonal to each other. In the same way, pilot signals(pilot symbols) 305 on channel A at time 6 on carrier 8 to carrier 11,pilot signals on channel B at time 6 on carrier 8 to carrier 11 andpilot signals on channel C at time 6 on carrier 8 to carrier 11 areorthogonal to each other.

Furthermore, pilot signals on channel A at time 12 on carrier 2 tocarrier 5, pilot signals on channel B at time 12 on carrier 2 to carrier5 and pilot signals on channel C at time 12 on carrier 2 to carrier 5are orthogonal to each other. In the same way, pilot signals on channelA at time 12 on carrier 8 to carrier 11, pilot signals on channel B attime 12 on carrier 8 to carrier 11 and pilot signals on channel C attime 12 on carrier 8 to carrier 11 are orthogonal to each other.

Furthermore, using, for example, the same sequence for pilot signals onchannel A at time 6 on carrier 2 to carrier 5 and for pilot signals onchannel C at time 6 on carrier 8 to carrier 11 and also using the samesequence for other pilot signals can reduce a circuit scale and obtaineffects similar to those in the case in FIG. 27. Here, a plurality ofconsecutive subcarriers have been explained as an example, but thisexample is not exclusive and similar effects can be obtained even ifpilot symbols 305 are assigned to subcarriers discretely to an extentthat orthogonality is not lost. Furthermore, as shown in FIG. 36, evenwhen pilot symbols are assigned on both the time axis and the frequencyaxis, similar effects can be obtained. In any case, effects similar tothose described above can be obtained by assigning pilot symbols 305 inthe frequency axis or the time domain in such a way that orthogonalityis not lost.

This embodiment has been explained using an example where pilot symbols305, which are orthogonal to each other in four-symbol units, but thepresent invention is not limited to four-symbol units. However,considering the influence of a fading fluctuation in the time directionor the frequency direction on the orthogonality, forming an orthogonalpattern with about 2 to 8 symbols may allow the estimation accuracy offrequency offset/phase noise to be secured. When the period of anorthogonal pattern is too long, the possibility that the orthogonalitymay not be secured increases and the estimation accuracy of frequencyoffset/phase noise degrades.

The important points in the method of configuration of pilot symbols 305of this embodiment are as follows:

(a) Pilot signals on the same carriers of channel A, channel B andchannel C are orthogonal to each other. (b) In different carriers onwhich pilot signals are arranged, channels using different sequencesexist. (c) Two or more channels using the same sequence exist (forexample, one certain sequence is used for both antenna A and antenna B,that is, one certain sequence is shared by a plurality of differentantennas).

In this way, when performing MIMO-OFDM transmission using threetransmission antennas, it is possible to minimize an increase oftransmission peak power without degrading the estimation accuracy offrequency offset/phase noise and realize a transmission apparatus in asimple configuration. It is an optimum condition to select pilot signalsso as to meet all of the three conditions above and form pilot carriers,but when, for example, only some of the above described effects arepreferably obtained, it is also possible to select pilot signals so atto meet, for example, only two out of the three conditions above andform pilot carriers.

This embodiment has been explained with an example using an OFDM schemebut the present invention is not limited to this and can also beimplemented in the same way when using a single-carrier scheme, othermulticarrier schemes or a spread spectrum communication scheme.Furthermore, this embodiment has been explained with an example wherethree antennas are used for transmission and reception respectively, butthe present invention is not limited to this. Embodiments with othernumbers of antennas or using other transmission methods will beexplained in detail later. In addition, this embodiment has beenexplained using naming such as “pilot symbol”, “reference symbol”,“guard symbol”, “preamble” here, but using other names will by no meansinfluence this embodiment. This will be the same in other embodiments.Furthermore, this embodiment has been explained using naming such as“channel A”, “channel B”, “channel C”, but these are used to facilitatethe explanations and using other names will by no means influence thisembodiment.

Embodiment 6

Embodiment 1 used the term “pattern” to explain the pilot symbolconfiguration, but this embodiment will use a term “sequence” as in thecase of Embodiment 5 to explain Embodiment 1. That is, this embodimentis similar to Embodiment 1 in the basic concept and the basicconfiguration.

FIG. 2 shows the configuration of MIMO-OFDM transmission apparatus 100according to this embodiment. However, FIG. 2 shows a case where thenumber of transmission antennas m=2 as an example.

Frame configuration signal generation section 112 receives controlinformation 111 on a modulation scheme or the like as input, generatesframe configuration signal 113 which includes information on the frameconfiguration and outputs this

Mapping section 102A receives transmission digital signal 101A ofchannel A, frame configuration signal 113 as input, generates basebandsignal 103A based on the frame configuration and outputs this.

Serial/parallel conversion section 104A receives baseband signal 103Aand frame configuration signal 113 as input, applies a serial/parallelconversion thereto based on frame configuration signal 113 and outputsparallel signal 105A.

Inverse Fourier transform section 106A receives parallel signal 105A asinput, applies an inverse Fourier transform thereto and outputs signal107A after the inverse Fourier transform.

Radio section 108A receives signal 107A after the inverse Fouriertransform as input, applies processing such as a frequency conversionand outputs transmission signal 109A. Transmission signal 109A is outputas a radio wave from antenna 110A.

MIMO-OFDM transmission apparatus 100 also generates transmission signal109B of channel B by applying processing similar to that on channel A tochannel B. The element indicated with “B” appended at the end of thereference numeral is the part related to channel B, which simply meansthat the target signal is not channel A but channel B and is basicallysubjected to processing similar to that on the above described partrelated to channel A indicated with “A” appended at the end of thereference numeral.

FIG. 3A shows an example of the configuration of a reception apparatusaccording to this embodiment. However, FIG. 3A shows a case where thenumber of reception antennas is n=2 as an example.

In reception apparatus 200, radio section 203X receives signal 202Xreceived at reception antenna 201X as input, applies processing such asa frequency conversion thereto and outputs baseband signal 204X.

Fourier transform section 205X receives baseband signal 204X as input,applies a Fourier transform a Id outputs signal 206X after the Fouriertransform.

A similar operation is carried out on the reception antenna 201Y sideand synchronization section 211 receives baseband signals 204X and 204Yas input, establishes time synchronization with a transmitter bydetecting, for example, reference symbols and outputs timing signal 212.The configuration or the like of reference symbols will be explained indetail using FIG. 4 or the like.

Frequency offset/phase noise estimation section 213 receives signals206X and 206Y after the Fourier transform as input, extracts pilotsymbols and estimates frequency offset and/or phase noise from the pilotsymbols and outputs phase distortion estimation signal 214 (phasedistortion including frequency offset). The configuration or the like ofpilot symbols will be explained in detail using FIG. 4 or the like.

Transmission path fluctuation estimation section 207A of channel Areceives signal 206X after the Fourier transform as input, extractsreference symbols of channel A, estimates a transmission pathfluctuation of channel A based on the reference symbols, for example,and outputs transmission path estimation signal 208A of channel A.

Transmission path fluctuation estimation section 207B of channel Breceives signal 206X after the Fourier transform as input, extractsreference symbols of channel B, estimates a transmission pathfluctuation of channel B based on the reference symbols, for example,and outputs transmission path estimation signal 208B of channel B.

Transmission path fluctuation estimation section 209A of channel A andtransmission path fluctuation estimation section 209B of channel Breceive a signal received at antenna 201Y as the target signal insteadof a signal received at antenna 201X and basically carry out processingsimilar to that described above on transmission path fluctuationestimation section 207A of channel A and transmission path fluctuationestimation section 207B of channel B.

Frequency offset/phase noise compensation section 215 receivestransmission path estimation signals 208A and 210A of channel A, andtransmission path estimation signals 208B and 210B of channel B, signals206X and 206Y after the Fourier transform and phase distortionestimation signal 214 as input, removes the phase the distortion of eachsignal and outputs transmission path estimation signals 220A and 222A ofchannel A after phase compensation, transmission path estimation signals220B and 222B of channel B after phase compensation and signals 221X and221Y after the Fourier transform and after the phase compensation.

Signal processing section 223 carries out, for example, an inversematrix calculation and outputs baseband signal 224A of channel A andbaseband signal 224B of channel B. More specifically, as shown in FIG.3B, for example, assuming that for a certain subcarrier, a transmissionsignal from antenna AN1 is Txa(t), a transmission signal from antennaAN2 is Txb(t), a received signal of antenna AN3 is R1(t), a receivedsignal of antenna AN4 is R2(t) and transmission path fluctuations areh11(t), h12(t), h21(t) and h22(t) respectively, the relationshipequation of Expression (1) holds.

Here, t is time, and n1(t) and n2(t) are noise. Signal processingsection 223 obtains a signal of channel A and a signal of channel B bycarrying out, for example, an operation of an inverse matrix usingEquation (1). Signal processing section 223 executes this operation onall subcarriers. h11(t), h12(t), h21(t) and h22(t) are estimated bytransmission path fluctuation estimation sections 207A, 209A, 207B and209B.

Frequency offset estimation/compensation section 225A receives basebandsignal 224A of channel A as input, extracts pilot symbols, estimates andcompensates for frequency offset of baseband signal 224A based on thepilot symbols and outputs baseband signal 226A after frequency offsetcompensation.

Channel A demodulation section 227A receives baseband signal 226A afterthe frequency offset compensation as input, demodulates data symbols andoutputs received data 228A.

MIMO-OFDM reception apparatus 200 also applies similar processing tobaseband signal 224B of channel B and obtains received data 228B.

FIG. 4 shows a time-frequency frame configuration of channel A (FIG.4(a)) and channel B (FIG. 4 (b)) of this embodiment. Signals at the sametime on the same carriers in FIG. 4(a) and FIG. 4 (b) are spatiallymultiplexed.

From time 1 to time 8, symbols for estimating transmission pathfluctuations corresponding to h11(t), h12(t), h21(t) and h22(t) inEquation (1), for example, symbols called “preambles” are transmitted.This preamble is composed of guard symbol 301 and reference symbol 302.Guard symbol 301 is assumed to be (0, 0) on the in-phase I-quadrature Qplane. Reference symbol 302 is, for example, a symbol at knowncoordinates other than (0, 0) on the in-phase I-quadrature Q plane.Channel A and channel B are configured such that no interference occurswith each other. That is, when, for example, guard symbol 301 is placedon channel A on carrier 1 at time 1, reference symbol 302 is placed onchannel B and when reference symbol 302 is placed on channel A oncarrier 2 at time 1, guard symbol 301 is placed on channel B, and inthis way different symbols are placed on channel A and channel B. Byadopting such an arrangement, when, for example, attention is focused onchannel A at time 1, it is possible to estimate a transmission pathfluctuation of carrier 3 using reference symbols 302 of carrier 2 andcarrier 4. Since carrier 2 and carrier 4 are reference symbols 302, thetransmission path fluctuation can be estimated. Therefore, transmissionpath fluctuations of all carriers of channel A can be accuratelyestimated at time 1. In the same way, transmission path fluctuations ofall carriers can be accurately estimated for channel B, too. From time 2to time 8, transmission path fluctuations of all carriers of channel Aand channel B can be estimated in the same way. Thus, since the frameconfiguration in FIG. 4 allows transmission path fluctuations of allcarriers to be estimated at all times from time 1 to time 8, this can besaid to be a preamble configuration capable of realizing quite highaccuracy estimation of transmission path fluctuations.

In FIG. 4, information symbol (data symbol) 303 is a symbol fortransmitting data. Here, suppose the modulation scheme is BPSK, QPSK,16QAM or 64QAM. A signal point constellation on the in-phaseI-quadrature Q plane or the like in this case will be explained indetail using FIG. 5.

Control symbol 304 is a symbol for transmitting control information on amodulation scheme, error correcting coding scheme, coding rate or thelike.

Pilot symbol 305 is a symbol for estimating a phase fluctuation byfrequency offset and/or phase noise. For example, a symbol having knowncoordinates on the in-phase I-quadrature Q plane is used As pilot symbol305. Pilot symbols 305 are placed on carrier 4 and carrier 9 on bothchannel A and channel B. When a wireless LAN builds a system at the samefrequency and in the same frequency band according to IEEE802.11a,IEEE802.11g and spatial multiplexing MIMO system, this allows the frameconfiguration to be shared, and therefore it is possible to simplify thereception apparatus.

FIG. 5 shows signal point constellations of BPSK, QPSK, 16QAM and 64QAMwhich are modulation schemes of information symbols 303 in FIG. 4 on thein-phase I-quadrature Q plane and normalization factors thereof.

FIG. 5A shows a signal point constellation of BPSK on the in-phaseI-quadrature Q plane and their coordinates are as shown in FIG. 5A. FIG.5B is a signal point constellation of QPSK on the in-phase I-quadratureQ plane and their coordinates are as shown in FIG. 5B. FIG. 5C is asignal point constellation of 16QAM on the in-phase I-quadrature Q planeand their coordinates are as shown in FIG. 5C. FIG. 5D is a signal pointconstellation of 64QAM on the in-phase I-quadrature Q plane and theircoordinates are as shown in FIG. 5D. FIG. 5E shows a relationshipbetween a modulation scheme and a multiplication factor (i.e.,normalization factor) for correcting signal point constellations in FIG.5A to FIG. 5D so that average transmit power is kept constant amongdifferent modulation schemes. For example, when transmission s carriedout under the modulation scheme of QPSK in FIG. 5B, as is clear fromFIG. 5E, it is necessary to multiply the coordinates in FIG. 5B by avalue of 1/sqrt(2). Here, sqrt(x) refers to the square root of x.

FIG. 6 shows an arrangement of pilot symbol 305 of FIG. 4 on thein-phase I-quadrature Q plane according to this embodiment. FIG. 6(a)shows an example of signal point constellation of pilot symbols 305 fromtime 11 to time 18 of carrier 4 of channel A shown in FIG. 4(a). FIG.6(b) shows an example of signal point constellation of pilot symbols 305from time 11 to time 18 on carrier 4 of channel B shown in FIG. 4(b).FIG. 6(c) shows an example of signal point constellation of pilotsymbols 305 from time 11 to time 18 of carrier 9 of channel A shown inFIG. 4(a). FIG. 6(d) shows an example of signal point constellation ofpilot symbols 305 from time 11 to time 18 on carrier 9 of channel Bshown in FIG. 4(b). Here, BPSK modulation is used for these arrangementsbut the modulation is not limited to this.

A feature of the signal point constellations of pilot symbols 305 inFIG. 6 is that the signal point constellations on the same carriers ofchannel A and channel B are orthogonal to each other (cross-correlationis zero).

For example, the signal point constellation from time 11 to time 14 ofchannel A on carrier 4 is orthogonal to the signal point constellationfrom time 11 to time 14 of channel B on carrier 4. Furthermore, the sameapplies to time 15 to time 18. The signal point constellation from time11 to time 14 of channel A on carrier 9 is also orthogonal to the signalpoint constellation from time 11 to time 14 of channel B on carrier 9.Furthermore, the same applies to time 15 to time 18. At this time, usinga Walsh-Hadamard conversion, orthogonal codes or the like is suitablebecause of the orthogonality of signals. FIG. 6 shows the case of BPSK,but if signals are orthogonal, QPSK modulation may also be used or itmay not be necessary to follow the rule of the modulation scheme.

Furthermore, in the case of this embodiment, for simplicity of thereceiver, suppose the signal point constellations are the same (samepattern) between carrier 4 of channel A and carrier 9 of channel B andbetween carrier 9 of channel A and carrier 4 of channel B. However, thesame pattern does not mean do adopt completely the same signal pointconstellation. For example, a case where only the phase relationship isdifferent on the in-phase I-quadrature Q plane can also be regarded asthe same pattern.

Here, as defined in Embodiment 5, if it is assumed that “the samesequence” means completely the same signal point constellation, forsimplicity of the receiver, the signal point constellations may beassumed to be same between carrier 4 of channel A and carrier 9 ofchannel B and between carrier 9 of channel A and carrier 4 of channel B,that is, the same sequence.

Furthermore, the signal point constellation of pilot symbols 305 is madeto differ between carriers 4 and 9 of channel A (or channel B), that is,different sequences, and this is because using the same signal pointconstellation may lead to an increase of transmission peak power.

Here, the advantage of orthogonality will be explained in detail usingFIG. 3A and FIG. 37 first.

FIG. 37 is an example of the configuration of frequency offset/phasenoise estimation section 213 in FIG. 3A. Pilot carrier extractionsection 602 receives signal 206X (or 206Y) after a Fourier transform asinput and extracts subcarrier which are pilot symbols 305. Morespecifically, it extracts signals of carrier 4 and carrier 9. Therefore,pilot carrier extraction section 602 outputs baseband signal 603 ofcarrier 4 and baseband signal 604 of carrier 9.

Sequence #1 storage section 3601 stores, for example, sequence #1 of “1,−1, 1, −1” in FIG. 6 and outputs signal 3602 of sequence 41 according totiming signal 212.

Sequence #2 storage section 3603 stores, for example, sequence #2 of “1,1, −1, −1” in FIG. 6 and outputs signal 3604 of sequence #2 according totiming signal 212.

Selection section 609 receives timing signal 212, signal 3602 ofsequence #1 and signal 3604 of sequence #2 as input, outputs the signalof sequence #2 as selection signal 610 and outputs the signal ofsequence #1 as selection signal 611.

Code multiplication section 612A receives baseband signal 603 of carrier4 and selection signal 611 as input, multiplies baseband signal 603 ofcarrier 4 by selection signal 611, generates baseband signal 613A ofchannel A of carrier 4 and outputs this. The reason is as follows.

Baseband signal 603 of carrier 4 is a signal in which the basebandsignal of channel A and the baseband signal of channel B aremultiplexed. On the other hand, when selection signal 611, that is, thesignal of sequence #1 is multiplied, the component of the basebandsignal of channel B whose cross-correlation is zero is removed and onlythe component of the baseband signal of channel A can thereby beextracted.

In the same way, code multiplication section 614A receives basebandsignal 604 of carrier 9 and selection signal 610 as input, multipliesbaseband signal 604 of carrier 9 by selection signal 610, generatesbaseband signal 615A of channel A of carrier 9 and outputs this.

Code multiplication section 612B receives baseband signal 603 of carrier4 and selection signal 610 as input, multiplies baseband signal 603 ofcarrier 4 by selection signal 610, generates baseband signal 613B ofchannel B of carrier 4 and outputs this.

Code multiplication section 614B receives baseband signal 604 of carrier9 and selection signal 611 as input, multiplies baseband signal 604 ofcarrier 9 by selection signal 611, generates baseband signal 615B ofchannel B of carrier 9 and outputs this.

As described above, by making signal point constellations of channel Aand channel B on the same carriers orthogonal to each other, even whenpilot symbols 305 are multiplexed on channel A and channel B, it ispossible to estimate frequency offset and/or phase noise with highaccuracy. Another important advantage is that since no channelestimation value (transmission path fluctuation estimation value) isrequired, it is possible to simplify the configuration of the part forcompensating for the frequency offset and/or phase noise. If signalpoint constellations of pilot symbols 305 of channel A and channel B arenot orthogonal to each other, signal processing of MIMO demultiplexing(for example, ZF, MMSE or MLD) is carried out, frequency offset and/orphase noise are then estimated. On the other hand, according to theconfiguration of this embodiment, it is possible to compensate forfrequency offset and/or phase noise before demultiplexing a signal(signal processing section 223) as shown in FIG. 3A. In addition, thefrequency offset and/or phase noise can be removed using pilot symbols305 even after demultiplexing the signal of channel A from the signal ofchannel B by signal processing section 223, thereby making it possibleto compensate for frequency offset and/or phase noise with higheraccuracy.

By the way, when the signal point constellations of channel A andchannel B of the same carriers are not orthogonal to each other, theestimation accuracy for frequency offset and/or phase noise by frequencyoffset/phase noise estimation section 213 in FIG. 3A decreases (signalsbecome components of interference with each other), and therefore it isdifficult to add the frequency offset/phase noise compensation section215 in FIG. 3A and it is not possible to realize high accuracy frequencyoffset/phase noise compensation.

Furthermore, this embodiment assumes the same signal point constellation(same sequence) for carrier 4 of channel A and carrier 9 of channel B,and carrier 9 of channel A and carrier 4 of channel B, thereby providingcommonality between sequence storage sections 3601 and 3603 in FIG. 37and this leads to simplification of the reception apparatus.

However, while it is essential in this embodiment that signal pointconstellations of channel A and channel B on the same carriers beorthogonal to each other, it is not necessarily essential that they havethe same sequence.

This embodiment has been explained using an example where pilot symbols305, which are orthogonal to each other in four symbol units, from time11 to time 14, but the present invention is not limited to four-symbolunits. However, when considering an influence on the orthogonality dueto a fluctuation of fading in the time direction, if an orthogonalpattern is formed in 2 to 8-symbol units, it may be possible to securethe estimation accuracy for frequency offset/phase noise. When theperiod of an orthogonal pattern is too long, the possibility that theorthogonality may not be secured increases and the estimation accuracyfor frequency offset/phase noise degrades.

FIG. 38 shows an example of the configuration of mapping section 102A(102B) of the transmission apparatus of this embodiment in FIG. 2. Datamodulation section 1103 receives transmission digital signal 101A (101B)and frame configuration signal 1102 as input, applies modulation totransmission digital signal 101A (101B) based on the information on themodulation scheme and timing included in frame configuration signal 1102and outputs modulated signal 1104 of data symbols 303.

Preamble mapping section 1105 receives frame configuration signal 1102as input and outputs modulated signal 1106 of preambles based on theframe configuration.

Sequence #1 storage section 3701 outputs signal 3702 of sequence #1. Inthe same way, sequence #2 storage section 3703 outputs signal 3704 ofsequence #2.

Pilot symbol mapping section 1111 receives signal 3702 of sequence #1,signal 3704 of sequence #2 and frame configuration signal 1102 as input,generates modulated signal 1112 of pilot symbol 305 and outputs this.

Signal generation section 1113 receives modulated signal 1104 of datasymbol 303, modulated signal 1106 of preambles and modulated signal 1112of pilot symbols 305 as input, generates baseband signal 103A (103B) inaccordance with the frame configuration and outputs this.

The above described explanation shows that adopting the configuration ofpilot symbols 305 as shown in FIG. 4 and FIG. 6 can simplify thereception apparatus. Similarly, adopting the pilot symbol configurationas shown in FIG. 4 and FIG. 6 also allows the transmission apparatus tostandardize sequence storage sections 3701 and 3703 as shown in FIG. 38and also leads to the simplification of the transmission apparatus.

As shown above, the method of generating preambles and pilot signals ofthis embodiment, the detailed configuration and operation of thetransmission apparatus which generates them and the reception apparatuswhich receives the modulated signal of this embodiment have beenexplained. According to this embodiment, it is possible to improve theestimation accuracy of frequency offset, transmission path fluctuationand synchronization, thereby improving the probability of detection ofsignals and simplifying the transmission apparatus and the receptionapparatus.

As in the case of Embodiment 1, the important points of the method ofconfiguration of pilot symbols 305 of this embodiment are as follows:

(a) Pilot signals of channel A and channel B on the same carriers areorthogonal to each other. (b) Within the same channel, differentsequences are used for different carriers on which pilot signals arearranged. (c) The same sequence is used on each channel (channel A andchannel B).

In this way, when performing MIMO-OFDM transmission using twotransmission antennas, it is possible to minimize an increase oftransmission peak power without degrading the estimation accuracy offrequency offset/phase noise and realize a transmission apparatus in asimple configuration. It is an optimum condition to select pilot signalsso as to meet all of the three conditions above and form pilot carriers.However, when, for example, only some of the above described effects arepreferably obtained, it is also possible to select pilot signals no atto meet, for example, only two out of the three conditions above andform pilot carriers.

This embodiment has been explained with an example using an OFDM schemebut the present invention is not limited to this and can also beimplemented in the same way when using single-carrier schemes, othermulticarrier schemes or spread spectrum communication schemes.Furthermore, this embodiment has been explained with an example wheretwo antennas are used for transmission and reception respectively, butthe present invention is not limited to this and can also be implementedeven when the number of reception antennas is three or more, in the sameway without being influenced. Furthermore, the frame configuration isnot limited to this embodiment and especially pilot symbols 305 forestimating distortion such as frequency offset, phase noise or the likemay only be required to be arranged on specific subcarriers, andtransmitted from a plurality of antennas, and the number of thesubcarriers for transmitting pilot symbols 305 is not limited to two ofthis embodiment. Embodiments with other numbers of antennas or usingother transmission methods will be explained in detail later. Inaddition, this embodiment has been explained using naming such as “pilotsymbol 305”, “reference symbol 302”, “guard symbol 301”, “preamble”, butusing other names will by no means influence this embodiment. This willbe the same in other embodiments. Furthermore, this embodiment has beenexplained using naming such as “channel A”, “channel B”, “channel C”,but these are used to facilitate the explanations and using other nameswill by no means influence this embodiment.

Furthermore, the frame configuration has been explained using the frameconfiguration in FIG. 4 as an example but this example is not exclusive.Especially, pilot symbols 305 have been explained with an example wherethey are arranged on specific subcarriers, but this example is notexclusive and the present invention can also be implemented in the sameway even when they are arranged as shown in FIG. 34, FIG. 35 and FIG. 36explained in Embodiment 5. However, it is important that they bearranged so as to secure orthogonality of pilot signals.

Embodiment 7

Embodiment 1 and Embodiment 6 have explained the method of arrangingpilot symbols 305 on two subcarriers in the case where the number oftransmission signals is two and the number of antennas is two, but thisembodiment will more specifically explain a method of arranging pilotsymbols 305 on four subcarriers and transmitting them.

FIG. 39 shows an example of the frame configuration of a transmissionsignal according to this embodiment and parts which correspond to thosein FIG. 4 are assigned the same reference numerals. FIG. 39(a) shows theframe configuration of channel A and FIG. 39(b) shows the frameconfiguration of channel B. As is clear from FIG. 39, pilot symbols(pilot carriers) 305 are arranged on carrier 3, carrier 5, carrier 8 andcarrier 10 except times at which reference symbols 302 and controlsymbols are transmitted.

FIG. 40 shows signal point constellations of pilot symbols 305 ofchannel A, channel B and a feature thereof. The feature at this time isthat the signal point constellations of channel A and channel B of thesame carriers are orthogonal to each other (cross-correlation is zero)as in the case of Embodiment 1.

For example, the signal point constellation (FIG. 40(a)) of channel A,carrier 3 from time 11 to time 14 and the signal point constellation(FIG. 40(b)) of channel B, carrier 3 from time 11 to time 14 areorthogonal to each other. Signal point constellations are designed insuch a way that orthogonality is also maintained from time 15 onward. Atthis time, using a Walsh-Hadamard conversion, orthogonal codes or thelike is suitable because of the orthogonality of signals. FIG. 40 showsa case of BPSK, but if signals are orthogonal, QPSK modulation may alsobe used or it may not be necessary to follow the rule of the modulationscheme.

Furthermore, for simplicity of the transmission apparatus and thereception apparatus, the same signal point constellation (same sequence)is used for carrier 3 of channel A (FIG. 40(a)) and carrier 10 ofchannel B (FIG. 40(h)), carrier 3 of channel B (FIG. 40(b)) and carrier5 of channel A (FIG. 40(c)), carrier 5 of channel B (FIG. 40(d)) andcarrier 8 of channel A (FIG. 40(e)), carrier 8 of channel B (FIG. 40(f))and carrier 10 of channel A (FIG. 40(g)). Here, the same sequence meanscompletely the same signal point constellation. Here, the respectivesequences are named sequence #1, sequence #2, sequence #3 and sequence#4 as shown in FIG. 40.

Furthermore, different signal point constellations (different sequences)of pilot symbols 305 are used for carriers 3, 5, 8, 10 of channel A andchannel B and this is because there is a possibility that using the samesignal point constellation (same sequence) may lead to an increase oftransmission peak power.

Here, the simplification of the transmission apparatus and the receptionapparatus and the necessity for orthogonality will be explained indetail using FIG. 41, FIG. 42 and FIG. 43.

FIG. 41 shows a configuration example of the MIMO-OFDM transmissionapparatus according to this embodiment. In FIG. 41, parts that operatein the same way as those in FIG. 2 and FIG. 29 are assigned the samereference numerals. MIMO-OFDM transmission apparatus 400 in FIG. 41 isdifferent from that in FIG. 29 in that there is no transmission sectionfor channel C but it operates in the same way as in FIG. 29 in otheraspects.

FIG. 42 shows an example of the detailed configuration of mappingsection 2802 in FIG. 41.

Data modulation section 2902 receives transmission data 2801 and frameconfiguration signal 113 as input, generates modulated signal 2903 ofdata symbol 303 according to frame configuration signal 113 and outputsthis.

Preamble mapping section 2904 receives frame configuration signal 113 asinput, generates modulated signal 2905 of preambles according to frameconfiguration signal 113 and outputs this.

Sequence #1 storage section 2906 outputs signal 2907 of sequence #1 inFIG. 40. Sequence #2 storage section 2908 outputs signal 2909 ofsequence #2 in FIG. 40. Sequence #3 storage section 2910 outputs signal2911 of sequence #3 in FIG. 40. Sequence #4 storage section 2912 outputssignal 2913 of sequence #4 in FIG. 40.

Pilot symbol mapping section 2918 receives signal 2907 of sequence #1,signal 2909 of sequence #2, signal 2911 of sequence #3, signal 2913 ofsequence #4 and frame configuration signal 113 as input, generatesmodulated signal 2919 of pilot symbol 305 according to frameconfiguration signal 113 and outputs this.

Signal generation section 2920 receives modulated signal 2903 of datasymbol 303, modulated signal 2905 of a preamble and modulated signal2919 of pilot symbol 305 as input and outputs modulated signal 103A ofchannel A and modulated signal 103B of channel B.

The configuration in FIG. 42 requires only four sequence storagesections. This is because as shown in FIG. 42, the present inventionuses a certain sequence for two or more subcarriers (used for twosubcarriers in FIG. 42). This allows the circuit scale of thetransmission apparatus to be reduced. On the other hand, unlike FIG. 42,when all different sequences are used, eight sequence storage sectionsare required and the circuit scale increases.

Next, the reception apparatus will be explained. FIG. 3 is an example ofthe configuration of the reception apparatus. Hereinafter, theconfiguration of frequency offset/phase noise estimation section 213 inFIG. 3 will be explained in detail using FIG. 43.

FIG. 43 is an example of the configuration of the frequency offset/phasenoise estimation section 213 in FIG. 3A according to this embodiment.Frequency offset/phase noise estimation section 213 in FIG. 43 isprovided with pilot carrier extraction section 3101, sequence storagesections 3108_1 to 3108_4 sequence selection section 3110, frequencyoffset/phase noise estimation section 3123_#3 of carrier 3, frequencyoffset/phase noise estimation section 3123_#5 of carrier 5, frequencyoffset/phase noise estimation section 3123_#8 of carrier 8 and frequencyoffset/phase noise estimation section 3123 #10 of carrier 10.

Pilot subcarrier extraction section 3101 receives signal 206X (or 206Y)after a Fourier transform as input and extracts subcarriers which arepilot symbols 305. More specifically, it extracts signals of carriers 3,5, 8 and 10. Therefore, pilot subcarrier extraction section 3101 outputsbaseband signal 3102_#3 of carrier 3, baseband signal 3102_#5 of carrier5, baseband signal 3102_#8 of carrier 8 and baseband signal 3102_#10 ofcarrier 10.

Sequence #1 storage section 3108_1 stores sequence #1 in FIG. 40 andoutputs signal 3109_1 of sequence #1 according to timing signal 212.Sequence #2 storage section 3108_2 stores sequence #2 in FIG. 40 andoutputs signal 3109_2 of sequence #2 according to timing signal 212.Sequence #3 storage section 3108_3 stores sequence #3 in FIG. 40 andoutputs signal 3109_3 of sequence #3 according to timing signal 212.Sequence #4 storage section 3108_4 stores sequence #4 in FIG. 40 andoutputs signal 3109_4 of sequence #4 according to timing signal 212.

Sequence selection section 3110 receives signal 3109_1 of sequence #1,signal 3109_2 of sequence #2, signal 3109_3 of sequence #3, signal3109_4 of sequence #4 and timing signal 212 as input and assignssequence #1 to signal 3111, sequence #2 to signal 3112, sequence #2 tosignal 3114, sequence #3 to signal 3115, sequence #3 to signal 3117,sequence #4 to signal 3118, sequence #4 to signal 3120, sequence 41 tosignal 3121 and outputs them.

Frequency offset/phase noise estimation section 3123_#3 of carrier 3 isprovided with code multiplication sections 3103A and 3103B, phasefluctuation estimation sections 3105A and 3105B and estimates frequencyoffset and/or phase noise of each channel of carrier 3.

Code multiplication section 3103A receives baseband signal 3102_#3 ofcarrier 3 and signal 3111 of sequence #1 as input and multipliesbaseband signal 3102_43 of carrier 3 by signal 3111 of sequence #1,thereby generating baseband signal 3104A_#3 of channel A of carrier 3and outputting this. The reason is as follows.

Baseband signal 3102_#3 of carrier 3 is a signal in which the basebandsignal of channel A and the baseband signal of channel B aremultiplexed. When this multiplexed signal is multiplied by signal 3111of sequence #1, the component of the baseband signal of channel B havinga zero cross-correlation is removed, thereby making it possible toextract only a baseband signal component of channel A.

Phase fluctuation estimation section 3105A receives baseband signal3104A 43 of channel A of carrier 3 as input, estimates a phasefluctuation based on this signal and outputs phase fluctuationestimation signal 3106A_43 of channel A.

In the same way, code multiplication section 3103B receives basebandsignal 3102_#3 of carrier 3 and signal 3112 of sequence #2 as input andmultiplies baseband signal 3102_43 of carrier 3 by signal 3112 ofsequence #2, thereby generating baseband signal 3104B_#3 of channel B ofcarrier 3 and outputting this.

Phase fluctuation estimation section 3105B receives baseband signal3104B #3 of channel B of carrier 3 as input, estimates a phasefluctuation based on this signal and outputs phase fluctuationestimation signal 3106B_#3 of channel B.

Frequency offset/phase noise estimation section 3123_#5 of carrier 5only differs in the signal to be processed and operates in the same wayas above described frequency offset/phase noise estimation section3123_#3 of carrier 3 and outputs phase fluctuation estimation signal3106A_#5 of channel. A about carrier 5 and phase fluctuation estimationsignal 3106B_#5 of channel B. Frequency offset/phase noise estimationsection 3123_#8 of carrier 8 also only differs in the signal to beprocessed and operates in the same way as above described frequencyoffset/phase noise estimation section 3123_#3 of carrier 3 and outputsphase fluctuation estimation signal 3106A_#8 of channel A about carrier8 and phase fluctuation estimation signal 3106B_#8 of channel B.Moreover, frequency offset/phase noise estimation section 3123_#10 ofcarrier 10 also only differs in the signal to be processed and operatesin the same way as above described frequency offset/phase noiseestimation section 3123_#3 of carrier 3 and outputs phase fluctuationestimation signal 3106A_#10 of channel A about carrier 10 and phasefluctuation estimation signal 3106B_#10 of channel B.

As described above, making signals of channel A and channel B of thesame carriers orthogonal to each other, allows frequency offset/phasenoise to be estimated with high accuracy even when pilot symbols 305 aremultiplexed on channel A and channel B. Another important advantage isthat since no channel estimation value (transmission path fluctuationestimation value) is required, the configuration of the part forcompensating for frequency offset and/or phase noise can be simplified.If the signal point constellations of channel A and channel B are notorthogonal to each other, signal processing of MIMO demultiplexing, forexample, ZF, MMSE, MLD is carried out and then frequency offset and/orphase noise are estimated. On the other hand, according to the presentinvention, it is possible to compensate for frequency offset and/orphase noise before demultiplexing a signal (signal processing section223) as shown in FIG. 3. In addition, signal processing section 223 canremove the frequency offset and/or phase noise using pilot symbols 305even after the signal is demultiplexed into the signal of channel A andsignal of channel B, and therefore it is possible to compensate for thefrequency offset and/or phase noise with higher accuracy.

On the other hand, when the signal point constellations of channel A andchannel B of the same carriers are not orthogonal to each other, theestimation accuracy for frequency offset/phase noise estimation section213 in FIG. 3A decreases (signals become components of interference witheach other), and therefore it is difficult to add frequency offset/phasenoise compensation section 215 in FIG. 3 and it is not possible toperform frequency offset/phase noise compensation with high accuracy.

As described above, a plurality of channels or a plurality of carriersare made to use pilot symbols 305 of the same sequence, and therefore itis possible to standardize sequence storage sections 3108_1 to 3108_4 inFIG. 43 among a plurality of channels or a plurality of carriers, whichleads to the simplification of the reception apparatus.

This embodiment has been explained using pilot symbols 305, which areorthogonal to each other in four-symbol units, as an example, but thepresent invention is not limited to four-symbol units. However,considering an influence of a fluctuation of fading in the tunedirection on the orthogonality, it may be possible to secure theestimation accuracy of frequency offset and/or phase noise if anorthogonal pattern is formed with about 2 to 8 symbols. When the periodof the orthogonal pattern is too long, the possibility that theorthogonality may not be secured increases and the estimation accuracyof frequency offset and/or phase noise degrades.

The important points of the method of configuration of pilot symbols 305of this embodiment are as follows.

(a) Pilot signals on the same carriers of channel A and channel B areorthogonal to each other. (b) For different carriers on which pilotsignals are arranged within the same channel, different sequences areused. (c) The same sequence is used for respective channels (channel Aand channel B).

In this way, when performing MIMO-OFDM transmission using twotransmission antennas, it is possible to minimize an increase oftransmission peak power without degrading the estimation accuracy offrequency offset and/or phase noise and realize a transmission apparatusin a simple configuration. It is an optimum condition to select pilotsignals so as to meet all of the three conditions above and form pilotcarriers, but when, for example, it is preferable to obtain only some ofthe above described effects, it is possible to select pilot signals soat to meet, for example, only two out of the above three conditions andform pilot carriers.

This embodiment has been explained with an example using an OFDM schemebut the present invention is not limited to this and can also beimplemented in the same way when using a single-carrier scheme, othermulticarrier schemes or a spread spectrum communication scheme.Furthermore, this embodiment has been explained with an example wheretwo antennas are used for transmission and reception respectively, butthe present invention is not limited to this and can also be implementedeven when the number of reception antennas is three or more in the sameway without being influenced. Furthermore, the frame configuration isnot limited to this embodiment and especially pilot symbols 305 forestimating distortion such as frequency offset, phase noise or the likemay only be required to be arranged on specific subcarriers, andtransmitted from a plurality of antennas, and the number of thesubcarriers for transmitting pilot symbols 305 is not limited to four inthis embodiment. Embodiments with other numbers of antennas or usingother transmission methods will be explained in detail later. Inaddition, this embodiment has been explained using naming such as “pilotsymbol 305”, “reference symbol 302”, “guard symbol 301”, “preamble”, butusing other names will by no means influence this embodiment. This willbe the same in other embodiments. Furthermore, this embodiment has beenexplained using naming such as “channel A” and “channel B”, but theseare used to facilitate the explanations and using other names will by nomeans influence this embodiment.

Furthermore, the frame configuration has been explained using the frameconfiguration in FIG. 39 as an example but is by no means limited tothis. Especially, pilot symbols 305 have been explained with the examplewhere they are arranged on specific subcarriers, but this example is notexclusive and the present invention can also be implemented in the sameway even when they are arranged as shown in FIG. 34, FIG. 35 and FIG. 36explained in Embodiment 5. However, it is important that they bearranged so as to secure orthogonality of pilot signals.

Embodiment 8

This embodiment will explain the method of configuration of preambles ina spatial multiplexing MIMO system in which the number of transmissionantennas is four and the number of transmission modulated signals isfour in detail.

FIG. 44 shows a configuration example of a MIMO-OFDM transmissionapparatus according to this embodiment. In FIG. 44, parts which operatein the same way as those in FIG. 2 are assigned the same referencenumerals as those in FIG. 2. In MIMO-OFDM transmission apparatus 4300,mapping section 4302 receives transmission digital signal 4301 and frameconfiguration signal 113 as input and outputs digital signal 103A ofchannel A, digital signal 103B of channel B, digital signal 103C ofchannel C and digital signal 103D of channel D.

The element indicated with “A” appended at the end of the referencenumeral is the part related to channel A, the element indicated with “B”appended is a part related to channel B, the element indicated with “C”appended is a part related to channel C and the element indicated with“D” appended is a part related to channel D, and they only differ insignals to be handled and are basically subjected to processing similarto processing on the part about channel A indicated with “A” appended atthe end of the reference numerals explained in Embodiment 1.

FIG. 45 shows an example of the frame configuration of a transmissionsignal formed by the transmission apparatus of this embodiment and partswhich correspond to those in FIG. 4 are assigned the same referencenumerals. FIG. 45(a) shows the frame configuration of channel A, FIG.45(b) shows the frame configuration of channel B, FIG. 45(c) shows theframe configuration of channel C and FIG. 45(d) shows the frameconfiguration of channel D.

In FIG. 45, the most characteristic aspect is the configuration ofpreambles and at a certain time on two channels, reference symbols 302and guard (null) symbols 301 are alternately arranged on the frequencyaxis (for example, channel A and channel C at time 1 and time 2 in FIG.45, and channel B and channel D at time 3 and time 4), while referencesymbols 302 are not arranged on the remaining two channels and onlyguard (null) symbols 301 are arranged (for example, channel B andchannel D at time 1 and time 2, and channel A and channel C at time 3and time 4 in FIG. 45).

When transmitting modulated signals of four channels, a method ofinserting reference symbol 302 for every 3 symbols may be thought of asthe simplest configuration, but it is also possible to think of a radiocommunication system in which a correlation of a transmission pathfluctuation on the frequency axis decreases due to the influence ofmultipaths as the distance on the frequency axis increases. In such aradio communication system, it is not preferable to insert referencesymbol 302 for every 3 symbols. But it is not always undesirable.Depending on a radio communication system, inserting reference symbol302 for every 3 symbols may not influence reception quality, either.

Therefore, this embodiment inserts reference symbols 302 and guard(null) symbols 301 alternately for OFDM symbols (generic name of symbolsof all subcarriers within a certain time, see FIG. 45(d)) at a certaintime. However, it is not possible to adopt the configuration in whichreference symbols 302 and guard (null) symbols 301 are arrangedalternately for OFDM symbols on all channels. This is because adoptingsuch a configuration would cause reference symbols 302 to collide witheach other among different channels. Even when the number of channelsincreases, adopting the frame configuration as shown in FIG. 45 canavoid reference symbols 302 from colliding with each other amongdifferent channels without reference symbols 302 being demultiplexedfrom each other too much on the frequency axis.

FIG. 46 shows a relationship between a modulation scheme of referencesymbols 302 and a normalization factor when four modulated signals aretransmitted.

FIG. 47 and FIG. 48 show examples of mapping when four modulated signalsare transmitted on the in-phase I-quadrature Q plane in OFDM symbolunits when the relationship between a modulation scheme of referencesymbols 302 and a normalization factor is as shown in FIG. 46. However,the examples in FIG. 47 and FIG. 48 apply to a case where thenormalization factor is 1. At this time, if the rule in Embodiment 3 isobserved, it is preferable to set the power of reference symbols 302 to2×2+0×0=4 as shown in FIG. 47 and FIG. 48. This allows the receiver toreduce the influence of quantization errors and thereby suppress areduction of reception quality.

Here, the case where the same average reception power is set for datasymbols 303 and preambles has been explained as an example, but thereare often cases where reception quality can be secured when the averagereception power of preambles is made to be greater than the averagereception power of data symbols 303. The above described concept canalso be applied to this case, too.

Furthermore, this embodiment has been explained using FIG. 45 as anexample of the transmission frame configuration in which it is possibleto prevent reference symbols 302 from colliding with each other amongdifferent channels without reference symbols 302 being demultiplexed toomuch from each other on the frequency axis even if the number ofchannels increases, but this example is not exclusive and it is alsopossible to obtain similar effects using a frame configuration as shownin FIG. 49.

Other Embodiments

The above described embodiments arrange symbols (preambles) forestimating a channel fluctuation at the head of a frame, but the presentinvention is not limited to this and symbols can be arranged at anypositions if it is possible to at least demultiplex data symbols 303.For example, it may be possible to use a method of inserting symbolsbetween data symbols 303 to improve the estimation accuracy or the like.

Furthermore, preambles are arranged on all carriers, that is, fromcarrier 1 to carrier 12 in FIG. 4 and FIG. 17 but preambles may also bearranged partially, for example, from carrier 3 to carrier 10.Furthermore, the above described embodiments use the term “preambles,”but this term itself does not have a specific meaning. The name is by nomeans a limiting one to this and may be reworded to, for example, “pilotsymbol control symbols.”

Furthermore, the present invention can be likewise applied to a casewhere one antenna illustrated in the above described embodiments is madeup of a plurality of antennas and a signal of the above described onechannel is transmitted using a plurality of antennas.

Moreover, the above described embodiments have used the term “channel”and this is one equation used to distinguish between signals transmittedfrom the respective antennas and replacing the term “channel” by a termlike a “stream” or “modulated signal” or “transmission antenna” or thelike will by no means influence the above described embodiments.

The invention claimed is:
 1. A transmission signal generation apparatuscomprising: a mapper configured to generate one or more transmissionsignals, each transmission signal including a data frame having preambleinformation, pilot information, and data information, wherein each ofthe one or more transmission signals includes: an associated preamblemultiplied by a factor so that an average reception power of theassociated preamble corresponds to an average reception power of thedata information received with the associated preamble, plural pilotsymbol sequences; and an Inverse Fourier transformer configured to:generate for each of the one or more transmission signals acorresponding Orthogonal Frequency Division Multiplexed (OFDM) signalfor transmission by a corresponding one of one or more antennas byInverse Fourier transforming each of the transmission signals, wherein:the Inverse Fourier transformer is configured to arrange: the pilotsymbol sequences in corresponding pilot carriers during a first timeperiod, and sets of the pilot carriers in a same carrier position in theOFDM signal, and wherein the plural pilot symbol sequences are allorthogonal to each other.
 2. The transmission signal generationapparatus of claim 1, wherein the one or more antennas includes one,two, three, or four antennas.
 3. The transmission signal generationapparatus of claim 1, wherein each transmission signal can includemultiple preamble sequences and multiple pilot subcarriers.
 4. Thetransmission signal generation apparatus of claim 1, wherein symbolvalues for the plural pilot symbol sequences are from a set {1, −1}. 5.The transmission signal generation apparatus of claim 1, wherein symbolvalues for the associated preambles are from a set {1+j, −1−j}.
 6. Atransmission signal generation method comprising: generating one or moretransmission signals, each transmission signal including a data framehaving preamble information, pilot information, and data information,wherein each of the one or more transmission signals includes: anassociated preamble multiplied by a factor so that an average receptionpower of the associated preamble corresponds to an average receptionpower of the data information received with the associated preamble, andplural pilot symbol sequences; performing Inverse Fourier transformoperations including: generating for each of the one or moretransmission signals a corresponding Orthogonal Frequency DivisionMultiplexed (OFDM) signal for transmission by a corresponding one of oneor more antennas by Inverse Fourier transforming each of thetransmission signals, arranging the pilot symbol sequences incorresponding pilot carriers during a first time period, and arrangingsets of the pilot carriers in a same carrier position in the OFDMsignal, wherein the plural pilot symbol sequences are all orthogonal toeach other.
 7. The transmission signal generation method of claim 6,wherein the one or more antennas includes one, two, three, or fourantennas.
 8. The transmission signal generation method of claim 6,wherein each transmission signal can include multiple preamble sequencesand multiple pilot subcarriers.
 9. The transmission signal generationmethod of claim 6, wherein symbol values for the plural pilot symbolsequences are from a set {1, −1}.
 10. The transmission signal generationmethod of claim 6, wherein symbol values for the plural pilot symbolsequences are from a set {1+j, −1−j}.
 11. A signal reception apparatuscomprising: processing circuitry including a Fourier transformerconfigured to input and process from one or more antennas one or moreOrthogonal Frequency Division Multiplexed (OFDM) signals, each OFDMsignal including a data frame having preamble information, pilotinformation, and data information, wherein each of the one or more OFDMsignals includes: an associated preamble multiplied by a factor so thatan average reception power of the associated preamble corresponds to anaverage reception power of the data information received with theassociated preamble, and plural pilot symbol sequences in correspondingpilot carriers during a first time period with sets of the pilotcarriers being in a same carrier position in the OFDM signal, whereinthe plural pilot symbol sequences are all orthogonal to each other. 12.The signal reception apparatus of claim 11, wherein the one or moreantennas includes one, two, three, or four antennas.
 13. The signalreception apparatus of claim 11, wherein each OFDM signal can includemultiple preamble sequences and multiple pilot subcarriers.
 14. Thesignal reception apparatus of claim 11, wherein symbol values for theplural pilot symbol sequences are from a set {1, −1}.
 15. The signalreception apparatus of claim 11, wherein symbol values for theassociated preambles are from a set {1+j, −1−j}.
 16. A signal receptionmethod comprising: inputting from one or more antennas and processingone or more Orthogonal Frequency Division Multiplexed (OFDM) signals,each OFDM signal including a data frame having preamble information,pilot information, and data information, wherein each of the one or moreOFDM signals includes: an associated preamble multiplied by a factor sothat an average reception power of the associated preamble correspondsto an average reception power of the data information received with theassociated preamble, and plural pilot symbol sequences in correspondingpilot carriers during a first time period with sets of the pilotcarriers being in a same carrier position in the OFDM signal, whereinthe plural pilot symbol sequences are all orthogonal to each other. 17.The signal reception method of claim 16, wherein the one or moreantennas includes one, two, three, or four antennas.
 18. The signalreception method of claim 16, wherein each transmission signal caninclude multiple preamble sequences and multiple pilot subcarriers. 19.The signal reception method of claim 16, wherein symbol values for theplural pilot symbol sequences are from a set {1, −1}.
 20. The signalreception method of claim 16, wherein symbol values for the plural pilotsymbol sequences are from a set {1+j, −1−j}.